Home Technology Adaptive fuzzy extended state observer for a class of nonlinear systems with output constraint
Article Open Access

Adaptive fuzzy extended state observer for a class of nonlinear systems with output constraint

  • Mahtab Delpasand and Mohammad Farrokhi EMAIL logo
Published/Copyright: November 28, 2023
Become an author with De Gruyter Brill

Abstract

In this study, an adaptive fuzzy extended state observer (AFESO) for single-input–single-output nonlinear affine systems in the presence of external disturbances and output constraints is proposed. In this regard, an extended state observer (ESO) was employed to estimate the unmeasured states and external disturbances simultaneously. To improve the estimation accuracy, the observer gains were adjusted using an adaptation law. To obtain a more comprehensive mathematical analysis and an accurate model for the ESO and to increase the degree of freedom, a Takagi–Sugeno fuzzy system was employed. The proposed AFESO relaxes the limitations of the ESO and improves the system performance as compared with the classical methods in the presence of time-varying disturbances. Next, a command-filtered backstepping controller is designed based on the barrier Lyapunov function method, which guarantees fast convergence of the tracking error as well as satisfying the output constraints of the system. The stability analysis showed that both the estimation error of the AFESO and the tracking error of the controller are bounded, and the tracking error converges to a small neighborhood of the origin. A simulating example of a flexible-joint manipulator shows the effectiveness of the proposed method as compared with the recently proposed method in the literature.

1 Introduction

The control of nonlinear systems in the presence of external disturbances and uncertainties is one of the crucial issues in control engineering and has been highly regarded by researchers in recent decades. Most systems are affected by external disturbances, which cause adverse effects on the system’s output and its performance [13]. Over the past decades, a variety of techniques have been proposed for disturbance rejection, including adaptive, intelligent, and robust control methods, all of which are considered anti-disturbance rejection control (ADRC) schemes [4]. Disturbance observer-based control [57], tracking differentiator [8,9], and extended state observer (ESO) [10,11] are some of the effective ADRC methods proposed in the literature. Furthermore, most controllers require full-state measurement of the dynamic systems to design controllers, which is not easy in practical applications, especially in the presence of uncertainties, disturbances, and constraints. Therefore, it is useful to design an observer such as ESOs to estimate unmeasured states and disturbances simultaneously.

ESOs are more common among other observers since the system is considered in the form of an integral-chain form. Moreover, ESOs require the least information in the design procedure, i.e., only the relative degree of the system is required [12]. This method can be applied to single-input–single-output high-order systems with matched disturbances [1315]. However, ESOs have limitations in the face of time-varying disturbances. That is, if time-varying disturbances with a wide range of changes are applied to the system, the ESO cannot eliminate the disturbances effectively. In the study of Gandhi and Adhyaru [16], the disturbance is considered to be constant or slow time-varying in order to eliminate its effects. In previous studies [17,18], higher-order disturbance derivatives are employed in the design of the generalized system to deal with this limitation and to eliminate the effects of time-varying disturbances at the cost of more computational time, complexity of the design procedure, and multiplicity of the design parameters. In the study of Naghdi and Sadrnia [19], to overcome this issue and increase flexibility in the observer gain design, a fuzzy linear ESO is designed to adjust the observer’s bandwidth. A time-varying ESO is designed in the study of Attar et al. [20] to enhance performance in the transient response. Nevertheless, since the ESO is linear with constant gains, the bandwidth of the observer is limited, which drives the system away from optimal performance and causes the peaking phenomenon. In the study of Yang et al. [21], a nonlinear extended state observer (NESO) with time-varying gain is designed to overcome the peaking phenomenon and eliminate the effect of time-varying disturbances.

In the design of ESOs, the observer gains play a crucial role in reducing the influence of the disturbance amplitude and improving the estimation performance [22]. Adjusting observer gains using an adaptive method is one of the effective approaches. In previous studies [23,24], an adaptive extended state observer (AESO) has been proposed to solve the problem of the ESO dependency on the bandwidth of the observer. This method reduces the bandwidth dependency by introducing an adaptation law to adjust the controller parameters. However, only the effect of the external disturbances is considered in the design procedure and not the effect of the state estimation error. In previous studies [25,26], an adaptive fuzzy observer is proposed to overcome this problem. In the study of Pyrkin et al. [27], an AESO is designed without knowledge of the boundary of the disturbances.

Another issue that affects a system’s performance is the constraints of the system that are encountered in practical systems and play crucial roles in the system’s stability. Constraint control problems can be divided into three categories: (i) output constraint [28,29], (ii) state constraint [30,31], and (iii) input constraint [32,33]. To deal with the constrained problems, the barrier Lyapunov functions (BLFs) have been proposed to ensure that constraints are not violated. In the study of Tran et al. [28], time-varying output constraints for robotic manipulators are considered by employing BLF and an adaptive controller. In this approach, the ESO is designed with linear and nonlinear terms that have significantly improved its performance in the presence of output constraints. In the study of Wang et al. [34], to guarantee the tracking error and avoid “complexity explosion” caused by the standard backstepping control method, the command-filtered backstepping controller is proposed. Compared with the standard backstepping controller [35], the command-filtered backstepping is designed to avoid the computational complexity problem caused by the repeated derivatives of the virtual control signals [36,37]. In the study of Cui et al. [38], to reduce the computational problem of the standard backstepping controller, a second-order filter is designed to process the virtual control signal. In previous studies [28,29], to prevent constraint violation, the boundary of the reference signal is considered considerably lower than the time-varying output constraints. In the study of Zhu et al. [39], an adaptive fuzzy observer is designed to estimate disturbances. To guarantee that the system’s output tracks the desired signal and the state constraints are not violated, a time-varying BLF is proposed.

Motivated by the aforementioned literature reviews, the objective of this study is to design an adaptive fuzzy observer to estimate the unmeasured states and the time-varying disturbances simultaneously using an ESO. In this regard, the Takagi–Sugeno (T–S) fuzzy system is employed to model nonlinear systems. It should be mentioned that using adaptive fuzzy observers is not a new idea. However, unlike other adaptive fuzzy observers, e.g., refs [19,4044], which employ Mamdani fuzzy systems to approximate the model, this study considers the T–S system to increase the degree of freedom in modeling and to obtain a more comprehensive mathematical analysis. In the convectional ESO, e.g., refs [20,45,46], nonlinearities, uncertainties, and unknown external disturbances are considered a lumped uncertainty that influences the accuracy of the estimation.

Compared with the convectional ESO, the proposed adaptive fuzzy extended state observer (AFESO) approximates the nonlinearities using fuzzy systems and estimates the external disturbances to improve the tracking accuracy and to obtain better performance of the closed-loop system. In traditional ESOs, e.g., refs [16,17,47,48], it is claimed that the time derivative of the disturbances converges to zero. Nevertheless, the estimation error can only converge to a compact set, whose boundaries depend on the upper bound of the disturbance derivatives. On the other hand, in this study, this dependency is relaxed in the ESO’s design and asymptotic stability of the estimation error dynamic is shown analytically in the presence of time-varying disturbances. In addition, the control objective is to design a control law to track the desired trajectory while ensuring all the closed-loop signals are bounded and the output constraint is not violated. To achieve these goals, the proposed controller is developed via the command-filtered backstepping method and the BLF strategy, which ensures that the output constraints are not violated and the estimation errors are bounded. In the constrained problems, to ensure that the output constraint is not violated, it is necessary to consider the constraints for the output estimation. Therefore, in this study, unlike refs [28,29,49], the estimation error is taken into account in the BLF to satisfy the output constraints. Moreover, as compared to previous studies [36,37], where the systems are considered in the form of strict feedback, in this study, a more general class of high-order nonlinear systems with output constraints is considered. To demonstrate the effectiveness of the proposed method, a simulation study is conducted on a single-link flexible-joint manipulator.

The main contributions of this study can be summarized as follows:

  1. A novel AFESO with adaptive gains is proposed to estimate the system states and to estimate the time-varying disturbances simultaneously with good accuracy.

  2. A combination of the BLF and the command-filtered backstepping controller is employed in the design procedure for the nth-order nonlinear systems to ensure less tracking errors as well as to consider the constraints of the system.

This article is organized as follows. The formulation of AFESO is introduced in Section 2. In Section 3, the proposed adaptive fuzzy observer is designed. The controller design is presented in Section 4. In Section 5, the stability analysis and disturbance rejection are presented. Section 6 presents the simulated results of a single-link flexible joint manipulator. Section 7 presents the conclusion.

2 Mathematical preliminaries

Consider the following nonlinear system with a relative degree of n:

(1) x ̇ ( t ) = f ( x ( t ) ) + g ( x ( t ) ) u ( t ) + b d d ( t ) ,

where x [ x 1 x n ] T R n × 1 is the vector of the system states, f ( x ( t ) ) R n and g ( x ( t ) ) R n are nonlinear vectors, u ( t ) is the input control signal, and d ( t ) R and b d R n are external disturbance and disturbance vector, respectively. Using matrix transformations, system (1) can be written in the following normal form:

(2) x ̇ i ( t ) = x i + 1 ( t ) i = 1 , .. . , n 1 x ̇ n ( t ) = f ( x ( t ) ) + g ( x ( t ) ) u ( t ) + d ( t ) y ( t ) = x 1 ( t ) ,

where f ( x ( t ) ) and g ( x ( t ) ) are smooth nonlinear functions and y ( t ) is the system’s output.

Assumption 1

It is assumed that the external disturbance d ( t ) is unknown but bounded and time varying. Moreover, its derivatives are bounded, i.e.,

(3) | d ( p ) ( t ) | Δ p p = 0 , 1 , 2 .

Assumption 2

It is assumed that the desired trajectory ( x d ) and its first-order derivative ( x ̇ d ) are known and bounded. Moreover, for any constant k c > 0 , there exists positive constants k y i > 0 ( i = 0 , 1 ) such that y < k c and x d ( i ) < k y i ( i = 0 , 1 ) .

In order to design a fuzzy observer, the nonlinear system in (2) can be modeled using a T–S system as follows:

(4) R j : if x 1 is G 1 j and and x n is G n j then x ̇ ( t ) = A j x ( t ) + b j u ( t ) + b d d ( t ) j = 1 , .. . , r y ( t ) = c j x ( t ) ,

where

A j 0 1 0 0 0 1 a 1 j a 2 j 0 0 a n j n × n , b j 0 0 1 n × 1 , b d 0 0 1 n × 1 , c j 1 0 0 1 × n ,

in which a i j ( , + ) are the known coefficients obtained from the linearization of f ( x ( t ) ) . Assuming that the states of the system are unmeasurable, an observer is required to estimate the states. Consequently, for the ESO design, system (4) can be converted into the following fuzzy extended system:

(5) R j : if x 1 is G 1 j and and x n is G n j then x ¯ ̇ ( t ) = A ¯ j x ¯ ( t ) + b ¯ j u ( t ) + E h ( t ) j = 1 , .. . , r y ¯ ( t ) = c ¯ j x ¯ ( t ) ,

where x ¯ [ x 1 x n + 1 ] T R ( n + 1 ) × 1 is the vector of the extended states, h ( t ) is the derivative of the disturbance, and system matrices are

A ¯ j A j ( n × n ) b d ( n × 1 ) 0 ( 1 × n ) 0 ( 1 × 1 ) ( n + 1 ) × ( n + 1 ) , b ¯ j b j ( n × 1 ) 0 ( 1 × 1 ) ( n + 1 ) × 1 , E 0 ( n × 1 ) 1 ( 1 × 1 ) ( n + 1 ) × 1 , c ¯ j c j ( n × 1 ) 0 ( 1 × 1 ) 1 × ( n + 1 ) .

Assumption 3

For j = 1 , .. . , r , the pairs ( A j , b j ) and ( A ¯ j , c ¯ j ) are controllable and observable, respectively.

Remark 1

A necessary condition for ( A ¯ j , c ¯ j ) being observable is that ( A j , c j ) is observable [17].

Remark 2

In order to design an observer, the fuzzy system must be observable. Therefore, in the process of the observer design, each fuzzy subsystem (and not the whole fuzzy T–S system) must be observable [50].

Proposition 1

For any constant k c > 0 , there exists positive constant k x > 0 such that the estimated state x ˆ 1 is bounded and satisfies x ˆ 1 < k x , which yields that the estimation error e 1 x 1 x ˆ 1 is also bounded and satisfies e 1 < k b , where k b k c k x .

3 Observer design

Since it is assumed that all system states are unmeasurable, an observer must be designed. To estimate the system states and the disturbance simultaneously, a fuzzy ESO is proposed as follows:

(6) R j : if x ˆ 1 is G 1 j and and x ˆ n is G n j then x ¯ ˆ ̇ ( t ) = A ¯ j x ¯ ˆ ( t ) + b ¯ j u ( t ) + l j ( t ) ( y ¯ ( t ) y ¯ ˆ ( t ) ) j = 1 , .. . , r y ¯ ˆ ( t ) = c ¯ j x ¯ ˆ ( t ) ,

where x ¯ ˆ [ x ˆ 1 x ˆ n + 1 ] T R ( n + 1 ) × 1 and y ¯ ˆ ( t ) are the estimation of the extended states and system’s output, respectively, A ¯ j , b ¯ j , and c ¯ j are the same as in (5), and l j ( t ) [ l 1 ( t ) l n + 1 ( t )] T is the time-varying observer gain that is adjusted by an adaptation law. The states and disturbance estimation errors are defined by

(7) e i x i x ˆ i , i = 1 , .. . , n + 1 .

Then, the estimation error dynamics for the jth subsystem can be calculated as

(8) e ̇ i ( t ) = e i + 1 ( t ) l i ( t ) e 1 ( t ) i = 1 , .. . , n 1 e ̇ n ( t ) = k = 1 n + 1 a k j e k ( t ) l n ( t ) e 1 ( t ) j = 1 , .. . , r e ̇ n + 1 ( t ) = h ( t ) l n + 1 ( t ) e 1 ( t ) ,

where a n + 1 j = 1 . The dynamics of the estimation error in (8) can be written as

(9) e ̇ ( t ) = A ˜ j ( t ) e ( t ) + b ˜ j h ( t ) , j = 1 , .. . , r ,

where e [ e 1 e n + 1 ] T R ( n + 1 ) × 1 is the estimation error and

A ˜ j l 1 ( t ) 1 0 0 l 2 ( t ) 0 1 0 0 a 1 j l n ( t ) a 2 j a n j 1 l n + 1 ( t ) 0 0 0 , b ˜ j 0 0 1 .

According to (9), it can be seen that the dynamics of the estimation error have been transformed into a time-varying system with an unknown input. Thus, the goal is to design an adaptive fuzzy ESO for the nonlinear system in (2), where the observer gains are adjusted using an adaptive law that guarantees stability of the estimation error in the presence of time-varying disturbances.

Lemma 1

[51] For all x R n × 1 and symmetric matrices A 1 and A 2 that satisfy x T A 1 x 0 and x T A 2 x > 0 , respectively, there is a positive constant τ that satisfies A 1 τ A 2 > 0 .

Theorem 1

Consider system (2) under Assumptions 1 and 3, Lemma 1, and Proposition 1. If the observer gains are adjusted by the following adaptation laws:

(10) l ̇ 1 = β 1 2 1 + β 1 2 e 1 2 e 1 2 ( l 1 2 l 2 l 1 ) + e 1 x ˆ 2 l 1 β 1 2 ( k b 2 e 1 2 ) l 1 k = 2 3 x ˆ k 2 l ̇ i = β i 2 1 + β i 2 e 1 2 e 1 2 ( l 1 l i l i + 1 l i ) l i k = 2 i + 2 x ˆ k 2 , i = 2 , .. . , n + 1 ,

and the following conditions are held, then the dynamics of the estimation error are asymptotically stable and the output constraint is satisfied:

(11) diag ( [ S 1 S 9 ] ) > 0 λ min ( M n + 1 ) > D 1 ,

where

S 1 γ 1 τ 1 μ 1 0.5 + 0.5 k b 2 e 1 2 0.5 + 0.5 k b 2 e 1 2 τ 1 μ 2 , S 2 γ 1 τ 2 μ 1 0.5 γ 1 0.5 γ 1 τ 2 μ 2 , S 3 γ n 1 τ 3 μ 3 0.5 γ n 1 a 1 0.5 γ n 1 a 1 τ 3 μ 4 ,

S 4 γ n 1 γ n 1 a n γ n 1 a n l n 1 2 , S 5 1 a n i = 1 n 1 a i l i a 1 l 1 + a 1 a n + γ n e 1 2 ( l 1 1 ) γ n k = 1 n + 1 x ˆ k 2 a 1 i = 1 n 1 a i l i a 1 2 a n + a 1 l n + 1 γ n l n + 1 e 1 2 ,

S 6 γ n γ n ( a j 1 a j + a j 2 a n ) γ n ( a j 1 a j + a j 2 a n ) l n 2 , S 7 γ n τ 4 μ 5 0.5 a 2 γ n 0.5 a 2 γ n τ 4 μ 6 γ n ( a 1 a 2 + a 2 2 a n ) , S 8 1 γ n + 1 e 1 4 γ n + 1 e 1 2 k = 1 n + 1 x ˆ k 2 γ n + 1 e 1 2 k = 1 n + 1 x ˆ k 2 1 γ n + 1 k = 1 n + 1 x ˆ k 2 2 , S 9 γ n + 1 0.5 γ n + 1 0.5 γ n + 1 l n + 1 2 ,

where γ i > 0 ( i = 1 , , n + 1 ) is the scaling factors; a i ( i = 1 , , n ) are the known elements of A j ; λ min ( M n + 1 ) is the smallest eigenvalue of M n + 1 ; τ i , μ i , and D 1 are positive constants; and M n + 1 is a positive definite matrix, which contains observer gains and scaling factors.

Proof. By considering the estimation error as e i x i x ˆ i ( i = 1 , .. . , n + 1 ) , the detailed procedures for proving the stability of estimation errors are described in the following steps.

Step 1. Consider the first state estimation error as e 1 x 1 x ˆ 1 . To guarantee that the output constraints are not violated, the BLF is proposed as follows:

(12) v 1 1 2 e 1 2 + 1 2 β 1 2 e ̇ 1 2 + 1 2 l 1 2 + 1 2 log k b 2 k b 2 e 1 2 ,

where e ̇ 1 is the derivative of e 1 and k b and β 1 are positive constants. The time derivative of v 1 is

(13) v ̇ 1 = e 1 e ̇ 1 + β 1 2 e ̇ 1 e ̈ 1 + l 1 l ̇ 1 + e 1 e ̇ 1 k b 2 e 1 2 ,

where e ̈ 1 is the second derivative of e 1 . Adding ± β 1 2 l 1 2 k = 1 3 e k 2 to (13) yields

(14) v ̇ 1 = e 1 e 2 e 3 T l 1 1 k b 2 e 1 2 + 1 + l 1 β 1 2 + β 1 2 ( l 1 2 l 2 l ̇ 1 ) 0 0 1 + β 1 2 ( l 2 2 l 1 2 + l ̇ 1 ) e 1 e 2 ( k b 2 e 1 2 ) l 1 β 1 2 + l 1 2 β 1 2 0 l 1 β 1 2 β 1 2 l 1 2 β 1 2 e 1 e 2 e 3 + l 1 l ̇ 1 + β 1 2 l 1 2 k = 2 3 ( x ˆ k 2 x k 2 + 2 x k e k ) + β 1 2 l 1 2 e 1 2 .

The adaptation law for l 1 is defined as

(15) l ̇ 1 β 1 2 1 + β 1 2 e 1 2 e 1 2 ( l 1 2 l 2 l 1 ) + e 1 x ˆ 2 l 1 β 1 2 ( k b 2 e 1 2 ) l 1 k = 2 3 x ˆ k 2 .

Eq. (14) can be written in the following compact form:

(16) v ̇ 1 = ε 1 T M 1 ε 1 ,

where ε 1 [ e 1 e 3 x 2 x 3 ] T is the augmented vector of the estimation errors and states of the system and

M 1 l 1 1 k b 2 e 1 2 + 1 + l 1 β 1 2 0 0 0 0 1 + β 1 2 ( l 2 2 l 1 2 + l ̇ 1 ) l 1 β 1 2 + l 1 2 β 1 2 0 0 0 l 1 β 1 2 β 1 2 l 1 2 β 1 2 0 0 1 k b 2 e 1 2 2 l 1 2 β 1 2 0 l 1 2 β 1 2 0 0 0 2 l 1 2 β 1 2 0 l 1 2 β 1 2 .

Hence, for the stability of the dynamic error, M 1 must be positive-definite. Therefore, the following condition must hold:

(17) l 1 l 1 β 1 2 + 1 + 1 k b 2 e 1 2 > 0 l 1 T S ¯ 1 l 1 > 0 ( l 1 + l 1 2 ) β 1 2 > 0 l 1 T S ¯ 2 l 1 > 0 ,

where l 1 = [ l 1 1 ] T and

S ¯ 1 γ 1 0.5 + 0.5 k b 2 e 1 2 0.5 + 0.5 k b 2 e 1 2 0 , S ¯ 2 γ 1 0.5 γ 1 0.5 γ 1 0

in which γ 1 β 1 2 . For all μ 1 , μ 2 > 0 , the following inequality holds:

(18) μ 1 l 1 2 + μ 2 > 0 l 1 T φ 1 l 1 > 0 ,

where φ 1 diag ( [ μ 1 , μ 2 ] ) . Based on Lemma 1, there exist positive constants τ 1 and τ 2 that satisfy S 1 S ¯ 1 τ 1 φ 1 > 0 and S 2 S ¯ 2 τ 2 φ 1 > 0 . Therefore, in order to satisfy conditions in (17), it suffices that the matrices S 1 and S 2 are positive-definite. Conditions in (17) can be written as the following linear matrix inequality (LMI):

(19) S 1 0 0 S 2 > 0 .

Step i. 2 i n 2 . A similar procedure as above is employed in this step. The i th state estimation error is e i x i x ˆ i . Then, the following Lyapunov function is considered:

(20) v i 1 2 e i 2 + 1 2 β i 2 e ̇ i 2 + 1 2 l i 2 ,

where e ̇ i is the derivative of e i and β i is a positive constant. The time derivative of v i is

(21) v ̇ i = e i e ̇ i + β i 2 e ̇ i e ̈ i + l i l ̇ i ,

where e ̈ i is the second derivative of e i . Adding ± β i 2 l i 2 k = 1 i + 1 e k 2 to (21) yields

(22) v ̇ i = e 1 e 2 e 3 e i 1 e i e i + 1 e i + 2 T β i 2 l i ( l i + l i + 1 + l ̇ i l 1 l i ) 0 0 0 0 β i 2 l i 2 β i 2 l i 2 0 0 0 0 0 β i 2 l i 2 0 0 0 0 0 0 0 0 0 l i 0 0 β i 2 l i 2 0 β i 2 ( l i + 1 + l ̇ i l 1 l i ) β i 2 l i 0 0 1 β i 2 l i 2 0 β i 2 l i 0 0 0 β i 2 β i 2 l i 2 e 1 e 2 e 3 e i 1 e i e i + 1 e i + 2 + β i 2 l i 2 k = 2 i + 1 ( x ˆ k 2 x k 2 + 2 x k e k ) + l ̇ i l i + β i 2 l i 2 e 1 2 .

The adaptation law for l i is defined as

(23) l ̇ i β i 2 1 + β i 2 e 1 2 e 1 2 ( l 1 l i l i + 1 l i ) l i k = 2 i + 2 x ˆ k 2 , i = 2 , .. . , n 2 .

Eq. (22) can be written in the following compact form:

(24) v ̇ i = ε i + 2 T M i ε i + 2 ,

where ε i + 2 [ e 1 e i + 2 x 2 x i + 2 ] T ( i = 2 , , n 2 ) and

M i β i 2 l i 2 0 0 0 0 0 β i 2 l i 2 β i 2 l i 2 0 0 0 0 0 0 β i 2 l i 2 0 0 0 0 0 0 0 0 0 0 0 0 0 l i 0 0 β i 2 l i 2 0 0 β i 2 ( l i + 1 + l ̇ i l 1 l i ) β i 2 l i 0 0 1 β i 2 l i 2 0 0 β i 2 l i 0 0 0 β i 2 β i 2 l i 2 0 0 2 β i 2 l i 2 0 0 0 0 0 0 2 β i 2 l i 2 0 0 0 0 β i 2 l i 2 .

M i β i 2 l i 2 0 0 0 0 0 β i 2 l i 2 β i 2 l i 2 0 0 0 0 0 0 β i 2 l i 2 0 0 0 0 0 0 0 0 0 0 0 0 0 l i 0 0 β i 2 l i 2 0 0 β i 2 ( l i + 1 + l ̇ i l 1 l i ) β i 2 l i 0 0 1 β i 2 l i 2 0 0 β i 2 l i 0 0 0 β i 2 β i 2 l i 2 0 0 2 β i 2 l i 2 0 0 0 0 0 0 2 β i 2 l i 2 0 0 0 0 β i 2 l i 2 .

Since M i is positive definite, then v ̇ i < 0 .

Step n −1. Similarly, the following Lyapunov function is considered for i = n 1 :

(25) v n 1 1 2 e n 1 2 + 1 2 β n 1 2 e ̇ n 1 2 + 1 2 l n 1 2 ,

where e ̇ n 1 is the derivative of e n 1 and β n 1 is a positive constant. The time derivative of v n 1 is

(26) v ̇ n 1 = e n 1 e ̇ n 1 + β n 1 2 e ̇ n 1 e ̈ n 1 + l n 1 l ̇ n 1 ,

where e ̈ n 1 is the second derivative of e n 1 . Adding ± β n 1 2 l n 1 2 k = 1 n + 1 e k 2 to (26) yields

(27) v ̇ n 1 = e T N 1 e + β n 1 2 l n 1 2 k = 2 n + 1 ( x ˆ k 2 x k 2 + 2 x k e k ) + l ̇ n 1 l n 1 β n 1 2 l n 1 ( l n + l ̇ n 1 l 1 l n 1 l n 1 ) e 1 2 ,

where e [ e 1 e n + 1 ] T is the estimation error vector and

N 1 β n 1 2 l n 1 ( l n 1 + a 1 ) 0 0 0 0 β n 1 2 l n 1 ( a 2 l n 1 ) β n 1 2 l n 1 2 0 0 0 β n 1 2 l n 1 a 3 0 β n 1 2 l n 1 2 0 0 0 0 0 0 0 0 β n 1 2 l n 1 a n 2 0 β n 1 2 l n 1 a n 1 + l n 1 0 0 0 β n 1 2 l n 1 2 0 β n 1 2 ( l n 1 a n a 1 + l n + l ̇ n 1 l 1 l n 1 ) β n 1 2 ( a 2 l n 1 ) β n 1 2 a 3 β n 1 2 a 4 β n 1 2 a n 1 1 β n 1 2 ( l n 1 2 a n ) 0 β n 1 2 l n 1 0 0 0 β n 1 2 β n 1 2 l n 1 2 .

The adaptation law for l n 1 is defined as

(28) l ̇ n 1 β n 1 2 1 + β n 1 2 e 1 2 e 1 2 ( l 1 l n 1 l n l n 1 ) l n 1 k = 2 n + 1 x ˆ k 2 .

Eq. (27) can be written in the following compact form:

(29) v ̇ n 1 = ε T M n 1 ε ,

where ε [ e 1 e n + 1 x 2 x n + 1 ] T and

M n 1 N 1 ( n + 1 ) × ( n + 1 ) 0 0 0 0 0 0 2 β n 1 2 l n 1 2 0 0 β n 1 2 l n 1 2 0 0 0 0 0 2 β n 1 2 l n 1 2 0 0 β n 1 2 l n 1 2 ( 2 n + 1 ) × ( 2 n + 1 ) .

For the stability of the dynamic error, M n 1 must be positive-definite. Therefore, the following condition must hold:

(30) β n 1 2 l n 1 ( l n 1 + a 1 ) > 0 l n 1 T S ¯ 3 l n 1 > 0 β n 1 2 ( l n 1 2 a n ) > 0 ,

where l n 1 = [ l n 1 1 ] T and

S ¯ 3 γ n 1 0.5 γ n 1 a 1 0.5 γ n 1 a 1 0 ,

where γ n 1 β n 1 2 . For all μ 3 , μ 4 > 0 , the following inequality is always maintained:

(31) μ 3 l n 1 2 + μ 4 > 0 l n 1 T φ 2 l n 1 > 0 ,

where φ 2 diag ( [ μ 3 , μ 4 ] ) . Based on Lemma 1, there exists a positive constant τ 3 that satisfies S 3 S ¯ 3 τ 3 φ 2 > 0 . Using the Schur complement theorem, conditions in (30) can be written as

(32) S 3 S ¯ 3 τ 3 φ 2 > 0 γ n 1 ( l n 1 2 a n ) > 0 S 4 γ n 1 γ n 1 a n γ n 1 a n l n 1 2 > 0

Hence, in order to establish the stability conditions in (30), it suffices that matrices S 3 and S 4 are positive-definite. Hence, the stability condition in (32) can be written as the following LMI:

(33) S 3 0 0 S 4 > 0 .

Step n . For i = n , the following Lyapunov function is considered:

(34) v n 1 2 e n 2 + 1 2 β n 2 e ̇ n 2 + 1 2 l n 2 ,

where e ̇ n is the derivative of e n and β n is a positive constant. The time derivative of v n is

(35) v ̇ n = e n e ̇ n + β n 2 e ̇ n e ̈ n + l n l ̇ n ,

where e ̈ n is the second derivative of e n . Adding ± β n 2 l n 2 k = 1 n + 1 e k 2 to (35) yields

(36) v ̇ n = e T N 2 e + β n 2 l n 2 k = 2 n + 1 ( x ˆ k 2 x k 2 + 2 x k e k ) + l n l ̇ n β n 2 l n ( l 1 l n l ̇ n l n + 1 l n ) e 1 2 ,

where e [ e 1 e n + 1 ] T is the vector of estimation error and

N 2 m 11 0 0 0 0 m 21 m 22 0 0 0 m 31 0 m 33 0 0 0 0 0 0 0 0 m ( n 1 ) 1 m ( n 1 ) 2 m ( n 1 ) 3 m ( n 1 ) ( n 1 ) 0 l n a 1 + m n 1 a 2 + m n 2 a 3 + m n 3 a n 1 + m n ( n 1 ) a n + m n n 0 m ( n + 1 ) 1 m ( n + 1 ) 2 m ( n + 1 ) 3 m ( n + 1 ) ( n 1 ) m ( n + 1 ) n m ( n + 1 ) ( n + 1 ) ,

m 11 = β n 2 l n 2 + ( a 1 l n ) i = 1 n a i l i a n a 1 2 a 1 ( l 1 l n l ̇ n l n + 1 ) + a 1 a n l n ,

m 21 = β n 2 a 2 i = 1 n a i l i a 1 2 2 a 1 a 2 a n a 2 ( l 1 l n l ̇ n l n + 1 ) + l n ( 2 a 1 + a 2 a n ) l n 2 m j 1 = β n 2 a j i = 1 n a i l i a 1 a j 1 2 a 1 a j a n a j ( l 1 l n l ̇ n l n + 1 ) + l n ( a j 1 + a j a n ) j = 3 , .. . , n + 1 ,

m j 2 = β n 2 ( l n a j a 1 a j a 2 a j 1 2 a 2 a j a n ) , m j j = β n 2 ( l n 2 a j 1 a j a j 2 a n ) j = 3 , .. . , n + 1 ,

m 22 = β n 2 ( l n 2 + l n a 2 a 1 a 2 a n a 2 2 ) ,

m j 3 = β n 2 ( a 3 a j 1 + a 2 a j + a 3 a j a n ) j = 4 , .. . , n + 1 ,

m ( n 1 ) j = β n 2 ( a j 2 a j 1 + 2 a n a n 1 a j + a n 1 a j 1 + a n 2 a j ) j = n 2 , .. . , n + 1 .

The adaptation law for l n is defined as

(37) l ̇ n β n 2 1 + β n 2 e 1 2 e 1 2 ( l 1 l n l n + 1 l n ) l n k = 2 n + 1 x ˆ k 2 .

Eq. (36) can be written in the following compact form:

(38) v ̇ n = ε T M n ε ,

where ε is the same as in (29) and

M n N 2 ( n + 1 ) × ( n + 1 ) 0 0 0 0 0 0 2 β n 1 2 l n 1 2 0 0 β n 1 2 l n 1 2 0 0 0 0 0 2 β n 1 2 l n 1 2 0 0 β n 1 2 l n 1 2 ( 2 n + 1 ) × ( 2 n + 1 ) .

For the stability of the dynamic error, M n must be positive-definite. Therefore, the following condition should hold:

(39) l n 2 + ( a 1 l n ) i = 1 n a i l i a n a 1 2 a 1 ( l 1 l n l ̇ n l n + 1 ) + a 1 a n l n > 0 β n 2 ( l n 2 a j 1 a j a j 2 a n ) > 0 j = 3 , .. . , n + 1 β n 2 ( l n 2 + l n a 2 a 1 a 2 a n a 2 2 ) > 0 .

Based on (37) and the Schur complement theorem, (39) can be written as

(40) l n T S 5 l n 1 > 0 S 5 1 a n i = 1 n 1 a i l i a 1 l 1 + a 1 a n + γ n e 1 2 ( l 1 1 ) γ n k = 1 n + 1 x ˆ k 2 a 1 i = 1 n 1 a i l i a 1 2 a n + a 1 l n + 1 γ n l n + 1 e 1 2 > 0 γ n ( l n 2 a j 1 a j a j 2 a n ) > 0 S 6 γ n γ n ( a j 1 a j + a j 2 a n ) γ n ( a j 1 a j + a j 2 a n ) l n 2 > 0 j = 3 , , n + 1 l n T S ¯ 7 l n 1 > 0 , S ¯ 7 γ n 0.5 a 2 γ n 0.5 a 2 γ n γ n ( a 1 a 2 + a 2 2 a n ) ,

where l n = [ l n 1 ] T and γ n β n 2 . For all μ 5 , μ 6 > 0 , the following inequalities always hold:

(41) μ 5 l n 2 + μ 6 > 0 l n T φ 3 l n > 0 ,

where φ 3 diag ( [ μ 5 , μ 6 ] ) . Based on Lemma 1, there exists a positive constant τ 4 that satisfies S 7 S ¯ 7 τ 4 φ 3 > 0 . Therefore, in order to establish the conditions in (39), it is sufficient that matrices S 5 , S 6 , and S 7 are positive-definite. The conditions in (39) can be written as the following LMI:

(42) S 5 0 0 0 S 6 0 0 0 S 7 > 0 .

Step n + 1. In the final design step, the Lyapunov function is defined as

(43) v n + 1 = 1 2 e n + 1 2 + 1 2 α n + 1 2 e ̇ n + 1 2 + 1 2 l n + 1 2 ,

where e ̇ n + 1 is the derivative of e n + 1 and β n + 1 is a positive constant. The time derivative of v n + 1 is

(44) v ̇ n + 1 = e n + 1 e ̇ n + 1 + β n + 1 2 e ̇ n + 1 e ̈ n + 1 + l n + 1 l ̇ n + 1 ,

where e ̈ n + 1 is the second derivative of e n + 1 . Adding ± β n 2 l n 2 k = 1 n + 1 e k 2 to (44) yields

(45) v ̇ n + 1 = e T N 3 e + d ̇ e n + 1 β n + 1 2 l n + 1 e 1 d ̈ + d ̈ d ̇ β n + 1 2 + β n + 1 2 ( l 1 l n + 1 l ̇ n + 1 ) e 1 d ̇ β n + 1 2 l n + 1 e 2 d ̇ + β n + 1 2 l n + 1 2 k = 2 n + 1 ( x ˆ k 2 x k 2 + 2 x k e k ) + l n l ̇ n e 1 2 β n + 1 2 ( l 1 l n + 1 2 l n + 1 2 ) ,

where e is the same as in (36). The adaptation law for l n + 1 is defined as

(46) l ̇ n + 1 β n + 1 2 1 + β n + 1 2 e 1 2 ( l 1 l n + 1 l n + 1 ) e 1 2 l n + 1 k = 2 n + 1 x ˆ k 2 .

Eq. (45) can be written as

(47) v ̇ n + 1 = ε T M 3 ε + d ̇ e n + 1 + d ̈ d ̇ β n + 1 2 β n + 1 2 l n + 1 e 1 d ̈ + β n + 1 2 ( l 1 l n + 1 l ̇ n + 1 ) e 1 d ̇ β n + 1 2 l n + 1 e 2 d ̇ ,

where ε is the same as in (29) and

M 3 N 3 ( n + 1 ) × ( n + 1 ) 0 0 0 0 0 0 2 β n 1 2 l n 1 2 0 0 β n 1 2 l n 1 2 0 0 0 0 0 2 β n 1 2 l n 1 2 0 0 β n 1 2 l n 1 2 ( 2 n + 1 ) × ( 2 n + 1 ) .

Using Young’s inequality and Assumption 1, (47) can be written as

(48) v ̇ n + 1 ε T M n + 1 ε + 1 2 1 β n + 1 2 + 3 β n + 1 2 d ̇ 2 + β n + 1 2 d ̈ 2 , ε T M n + 1 ε + D 1

where D 1 1 2 1 β n + 1 2 + 3 β n + 1 2 Δ 1 2 + β n + 1 2 Δ 2 2 is a positive constant, Δ i is the same as in (3), and matrix M n + 1 is

M n + 1 1 2 β n + 1 2 ( l n + 1 2 l 1 2 l n + 1 2 l ̇ n + 1 2 ) 0 0 0 0 0 0 β n + 1 2 l n + 1 2 1 2 β n + 1 2 l n + 1 2 0 0 0 0 0 0 β n + 1 2 l n + 1 2 0 0 0 l n + 1 0 0 β n + 1 2 l n + 1 2 1 2 0 0 2 β n + 1 2 l n + 1 2 0 0 β n + 1 2 l n + 1 2 0 0 2 β n + 1 2 l n + 1 2 0 0 0 β n + 1 2 l n + 1 2 .

For the stability of the dynamic error, M n + 1 must be positive-definite. Therefore, the following conditions should hold:

(49) l n + 1 2 l 1 2 l n + 1 2 l ̇ n + 1 2 > 0 β n + 1 2 l n + 1 2 1 2 > 0 λ min ( M n + 1 ) > D 1 .

Based on (46) and the Schur complement theorem, (49) can be written as

(50) l 1 T S 8 l 1 > 0 S 8 1 γ n + 1 e 1 4 γ n + 1 e 1 2 k = 1 n + 1 x ˆ k 2 γ n + 1 e 1 2 k = 1 n + 1 x ˆ k 2 1 γ n + 1 k = 1 n + 1 x ˆ k 2 2 > 0 γ n + 1 l n + 1 2 1 2 > 0 S 9 γ n + 1 0.5 γ n + 1 0.5 γ n + 1 l n + 1 2 > 0 λ min ( M n + 1 ) > D 1 ,

where γ n + 1 β n + 1 2 and l 1 is the same as in (17). Therefore, in order to establish the conditions in (49), it is sufficient that matrices S 8 and S 9 are positive definite. The conditions in (49) can be written as the following LMI:

(51) S 8 0 0 S 9 > 0 λ min ( M n + 1 ) > D 1 .

Therefore, according to the following adaptation laws:

(52) l ̇ 1 = β 1 2 1 + β 1 2 e 1 2 e 1 2 ( l 1 2 l 2 l 1 ) + e 1 x ˆ 2 l 1 β 1 2 ( k b 2 e 1 2 ) l 1 k = 2 3 x ˆ k 2 l ̇ i = β i 2 1 + β i 2 e 1 2 e 1 2 ( l 1 l i l i + 1 l i ) l i k = 2 i + 2 x ˆ k 2 , i = 2 , .. . , n + 1

and satisfaction of the following conditions:

(53) diag ( [ S 1 S 9 ] ) > 0 λ min ( M n + 1 ) > D 1 ,

the dynamics of the estimation error are asymptotically stable (i.e., v ̇ 1 + + v ̇ n + 1 < 0 ).

Remark 3

Conditions (53) are not restrictive. By properly adjusting the initial conditions of the observer gain and the scaling parameters ( β i ), these conditions can always be satisfied.

4 Controller design

The objective is to design a control law ( u ) to ensure that the system output ( y = x 1 ) asymptotically tracks the desired trajectory ( x d ). To eliminate the effects of the external disturbances and stabilization of the closed-loop system, the proposed control is designed based on the command-filterer backstepping controller. In addition, to avoid violation of the output constraint, the BLF is employed. To design a command-filtered backstepping with BLF, the tracking errors are defined as follows:

(54) z 1 x 1 x 1 , c z i x ˆ i x i , c i = 2 , .. . , n + 1 ,

where x i , c represents the output variable of the filters and x 1 , c = x d . The command-filtered backstepping is designed to reduce conservatism. That is, the higher-order derivatives of the desired trajectory need not be bounded, available, and known.

Lemma 2

[52] A command filter with a unity DC gain is given by

(55) x ̇ i , c = ω i ( x i , c α i 1 ) , i = 2 , , n + 1 ,

where ω i > 0 is the filter bandwidth, and α i 1 and x i , c are the input and output variables of the command filter, respectively. The initial condition is selected as x i , c ( 0 ) = α i 1 ( z i 1 ( 0 ) , x ̇ i 1 , c ( 0 ) ) , which yields x ̇ i , c ( 0 ) = 0 .

Remark 4

In (55), a first-order low-pass filter is designed to simplify the analysis. However, higher-order filters can also be designed. For instance, a second-order filter is discussed in the study of Wang and Yu [36].

The compensated tracking errors are introduced as

(56) ϑ i z i ξ i , i = 1 , .. . , n + 1 .

To ensure that the error compensation system can compensate for the error, the following error compensation system with the initial condition ξ i ( 0 ) = 0 is given as follows:

(57) ξ ̇ i k i ξ i + θ i ( ξ i + 1 + x i + 1 , c ) α i + χ i ϑ i 1 i = 1 , .. . , n + 1 ,

where

χ i 1 k v 2 ϑ 1 2 i = 2 1 i = 3 , .. . , n + 1 , θ i 1 i = 1 , .. . , n 1 0 i = n , n + 1

in which k i > 0 are design parameters, k v > 0 , and α i are the virtual control signals of the filter that are defined as

(58) α i k i z i + x ̇ i , c ϒ i ( e ) , i = 1 , , n + 1 ,

where

ϒ i ( e ) e ̇ 1 + l 1 e 1 i = 1 l i e 1 i = 2 , .. . , n + 1 .

The control law is divided into two parts: u s j and u f j . The control u s j is the output feedback for system stabilization and u f j is designed to eliminate the tracking error. Therefore, the control law is designed for the jth fuzzy subsystem as follows:

(59) u j = 1 b j ( u s j + u f j ) ,

where

(60) u s j i = 1 n + 1 κ i x ˆ i ,

(61) u f j α u = i = 1 n + 1 ( a i κ i ) ( ξ i + x i , c ) ,

where κ n + 1 R is the disturbance compensation gain and κ i [ κ 1 .. . κ n ] R 1 × n is the feedback control gain vector and is designed such that A f j A j b j κ j is Hurwitz. The disturbance compensation gain is designed as [17]:

(62) κ n + 1 [ c j ( A j b j κ j ) 1 b j ] 1 c j ( A j b j κ j ) 1 b d .

5 Stability analysis of the closed-loop system

In this section, stability analysis is established to confirm that all signals in the closed-loop system are ultimately bounded, asymptotic output tracking is achieved, and output constraint is not violated. Therefore, to illustrate the stability of the closed-loop system, the detailed design procedures of the compensated tracking error dynamics are described in the following steps.

Step 1. Using z 1 x 1 x 1 , c , the derivative of the compensated tracking error ϑ 1 z 1 ξ 1 is

(63) ϑ ̇ 1 = x 2 x ̇ 1 , c ξ ̇ 1 .

Since the states of the system cannot be measured, (63) is rewritten as

(64) ϑ ̇ 1 = e 2 + x ˆ 2 x ̇ 1 , c ξ ̇ 1 = e 2 + z 2 + x 2 , c x ̇ 1 , c ξ ̇ 1 .

Substituting (55) and (57) into (64) yields

(65) ϑ ̇ 1 = k 1 ϑ 1 + ϑ 2 .

Step i , 2 i n 1 . Using a similar procedure as in step 1 and using z i x ˆ i x i , c , the derivative of the compensated tracking error ϑ i z i ξ i is

(66) ϑ ̇ i = x ˆ ̇ i x ̇ i , c ξ ̇ i .

Substituting (55) and (57) into (66) yields

(67) ϑ ̇ i = ϑ i 1 k i ϑ i + ϑ i + 1 .

Step n . By defining ϑ n z n ξ n , the time derivative of ϑ n is

(68) ϑ ̇ n = i = 1 n + 1 a i j x ˆ i + b j u j + l n e 1 x ̇ n , c ξ ̇ n j = 1 , .. . , r ,

where a n + 1 j = 1 . Substituting (59) into (68) yields

(69) ϑ ̇ n = i = 1 n + 1 a i j x ˆ i + u s j + u f j + l n e 1 x ̇ n , c ξ ̇ n = i = 1 n + 1 ( a i j κ i ) x ˆ i + α u + l n e 1 x ̇ n , c ξ ̇ n = ( a 1 j κ 1 ) e 1 + i = 1 n + 1 ( a i j κ i ) ( ϑ i + ξ i + x i , c ) + α u ϑ n 1 k n ϑ n = ( a 1 j κ 1 ) e 1 + i = 1 n + 1 ( a i j κ i ) ϑ i ϑ n 1 k n ϑ n .

Step n + 1. By defining ϑ n + 1 z n + 1 ξ n + 1 , the time derivative of ϑ n + 1 is

(70) ϑ ̇ n + 1 = l n + 1 e 1 x ̇ n + 1 , c ξ ̇ n + 1 = k n + 1 ϑ n + 1 ϑ n .

According to (65), (67). (69), and (70), the dynamics of the compensated tracking errors can be calculated as

(71) ϑ ̇ 1 = k 1 ϑ 1 + ϑ 2 ϑ ̇ i = ϑ i 1 k i ϑ i + ϑ i + 1 i = 2 , .. . , n 1 ϑ ̇ n = ( a 1 j κ 1 ) e 1 + i = 1 n + 1 ( a i j κ i ) ϑ i ϑ n 1 k n ϑ n j = 1 , .. . , r ϑ ̇ n + 1 = k n + 1 ϑ n + 1 ϑ n .

The stability of the compensated tracking error dynamics (71) is shown in Theorem 2.

Lemma 3

[53] For all ϑ 1 that satisfies ϑ 1 < k v with any constant k v > 0 ,

log k v 2 k v 2 ϑ 1 2 ϑ 1 2 k v 2 ϑ 1 2 .

holds, where k v k c k y 0 .

Theorem 2

Consider system (2) under Assumptions 1–3 and Proposition 1 with the output feedback controller given by (59), the command-filtered backstepping in (55), the compensated tracking error signals in (56), and the virtual control signal in (58). Then, all signals of the closed-loop system are bounded, the output constraint is not violated, and the system (2) is exponentially stable until V T σ / Ψ , where V T is the Lyapunov function and Ψ and σ are positive constants.

Proof. To illustrate the stability of the closed-loop system, the Lyapunov function is defined as

(72) V T V 1 + V 2 + + V n + 1 ,

where V 1 1 2 log k v 2 k v 2 ϑ 1 2 is a BLF function, whose time derivative can be calculated as

(73) V ̇ 1 = ϑ 1 ϑ ̇ 1 k v 2 ϑ 1 2 = k 1 ϑ 1 2 + k z ϑ 1 ϑ 2 ,

where k z 1 / ( k v 2 ϑ 1 2 ) and k 1 k 1 k z . For 2 i n 1 , considering the Lyapunov candidate as V i ( 1 / 2 ) ϑ i 2 , the time derivative of V i yields

(74) V ̇ i = ϑ i ϑ ̇ i = ϑ i 1 ϑ i k i ϑ i 2 + ϑ i ϑ i + 1 , 2 i n 1 .

Considering V n 1 2 ϑ n 2 as the nth Lyapunov candidate, the time derivative of V n is

(75) V ̇ n = ϑ n ϑ ̇ n = ϑ n ( a 1 j κ 1 ) e 1 + i = 1 n + 1 ( a i j κ i ) ϑ i ϑ n 1 k n ϑ n .

Based on Young’s inequality, (75) is rewritten as

(76) V ̇ n 1 2 ( a 1 j κ 1 ) 2 e 1 2 + 1 2 i = 1 n + 1 ( a i j κ i ) 2 ϑ i 2 + n + 2 2 k n ϑ n 2 ϑ n 1 ϑ n .

Considering V n + 1 1 2 ϑ n + 1 2 as the (n + 1)th Lyapunov function, the time derivative of V n + 1 is

(77) V ̇ n + 1 = ϑ n + 1 ϑ ̇ n + 1 = ϑ n + 1 ( k n + 1 ϑ n + 1 ϑ n ) .

Based on (72), the time derivative of V T is

(78) V ̇ T = V ̇ 1 + V ̇ 2 + + V ̇ n + 1 i = 1 n + 1 k i ϑ i 2 + 1 2 ( a 1 j κ 1 ) 2 e 1 2 + 1 2 i = 1 n + 1 ( a i j κ i ) 2 ϑ i 2 + n + 2 2 k n ϑ n 2 1 2 i = 1 n + 1 ( 2 k i ( a i j κ i ) 2 ) ϑ i 2 + n + 2 2 k n ϑ n 2 + 1 2 ( a 1 j κ 1 ) 2 k b 2 1 2 i = 1 n + 1 ς i ϑ i 2 + σ ,

where

ς i 2 k i ( a i j κ i ) 2 n + 2 2 + k n i = n 2 k i ( a i j κ i ) 2 otherwise ,

σ 1 2 ( a 1 j κ 1 ) 2 k b 2 .

Using Lemma 3, (78) is written as

(79) V ̇ T ς 1 2 log k v 2 k v 2 ϑ 1 2 1 2 i = 2 n + 1 ς i ϑ i 2 + σ Ψ V T + σ ,

where Ψ = min { ς i } ( i = 1 , .. . , n + 1 ) . The compensated tracking error is stable if κ i and k i are selected such that Ψ is positive. Therefore, system (2) is exponentially stable until V T σ Ψ , and the parameters of the controller are bounded.

Remark 5

It is important to note that for better performance of the command-filtered backstepping, the filter bandwidth ( ω i ) is selected much greater than k i .

Corollary 1

According to Theorem 2 and the Lyapunov function in (72) , the compensated tracking error remains within the compact set Ω v = ϑ i ϑ i 2 V T ( 0 ) e Ψ t + 2 σ Ψ + k v 2 .

Proof. Multiplying both sides of (79) by e Ψ t and integrating over the time range of [ 0 , t ] yields

(80) V T V T ( 0 ) σ Ψ e Ψ t + σ Ψ V T ( 0 ) e Ψ t + σ Ψ .

Using (72) yields

(81) 1 2 i = 2 n + 1 ϑ i 2 V T V T ( 0 ) e Ψ t + σ Ψ .

Adding 1 2 ν 1 2 to both sides of (81) gives

(82) 1 2 i = 1 n + 1 ϑ i 2 V T ( 0 ) e Ψ t + σ Ψ + 1 2 ϑ 1 2 .

Based on Lemma 3, (82) becomes

(83) 1 2 i = 1 n + 1 ϑ i 2 V T ( 0 ) e Ψ t + σ Ψ + 1 2 k v 2 .

Inequality (83) implies that

(84) ϑ i 2 2 V T ( 0 ) e Ψ t + 2 σ Ψ + k v 2 .

Therefore, it is straightforward to verify that ϑ i converges to the compact set

Ω v = ϑ i ϑ i 2 V T ( 0 ) e Ψ t + 2 σ Ψ + k v 2 .

In addition, ϑ i can be made arbitrarily small by selecting appropriate values for the design parameters.

Corollary 2

Consider Theorems 1 and 2. The adaptation laws (52) are stable and converge into the constant values of lim t l i ( t ) = l i 0 .

Proof. According to Theorem 1 and the Lyapunov stability theory, the stability of estimation errors is guaranteed. That is,

(85) lim t e i ( t ) = 0 lim t ( x i ( t ) x ˆ i ( t ) ) = 0 lim t x ˆ i ( t ) = x i ( t ) .

Taking the limits of the adaptation laws in (52) yields

(86) lim t l ̇ i ( t ) = lim t β 1 2 1 + β 1 2 e 1 2 e 1 2 ( l 1 2 l 2 l 1 ) + e 1 x ˆ 2 l 1 β 1 2 ( k b 2 e 1 2 ) l 1 k = 2 3 x ˆ k 2 lim t β i 2 1 + β i 2 e 1 2 e 1 2 ( l 1 l i l i + 1 l i ) l i k = 2 i + 2 x ˆ k 2 i = 2 , .. . , n + 1 .

Based on (85) and Theorem 2, and because all signals in the closed-loop systems are exponentially stable, it yields

(87) lim t l ̇ i ( t ) = 0 .

Additionally, due to the adaptive laws in (52), the convergence of the adaptive parameters is ensured. That is,

lim t e i ( t ) = 0 lim t l ̇ i ( t ) = 0 lim t l i ( t ) = l i 0 ,

where l i 0 is a positive constant.

The proposed approach can be summarized as the following algorithm:

Step 1: Model the nonlinear system in (1) using the T–S linear fuzzy system in (4).

Step 2: Convert system (4) into the extended fuzzy system in (5).

Step 3: Design fuzzy ESO in (6) and adjust the observer gains using the adaptation laws in (52) for each subsystem.

Step 4: Adjust the scaling parameters and the initial conditions to satisfy conditions in (53).

Step 5: Design the controller using (59), the command-filtered backstepping in (55), the compensated tracking error signals in (56), and the virtual control signal in (58).

The block diagram of the proposed observer in the presence of external disturbance is shown in Figure 1. The controller is synthesized via BLF and the command-filtered backstepping method using output feedback.

Figure 1 
               Block diagram of the proposed AFESO method.
Figure 1

Block diagram of the proposed AFESO method.

6 Simulating example

To verify the effectiveness of the proposed observer and controller, a single-link flexible-joint manipulator is simulated in the presence of external disturbances and measurement noise. The simulation results are compared with the NESO methods in the study of Tran et al. [28] and the ESO scheme in the study of Rsetam et al. [54], in the presence of output constraints.

The nonlinear model of a single-link flexible-joint manipulator in the presence of external disturbance can be written as [55]

(88) I q ̈ 1 ( t ) + M g L sin ( q 1 ( t ) ) + K ( q 1 ( t ) q 2 ( t ) ) = 0 J q ̈ 2 ( t ) K ( q 1 ( t ) q 2 ( t ) ) = u ( t ) + d ( t ) ,

where q 1 and q 2 are the angular displacement of the link and the motor in rad., respectively, q 1 q 2 is the elastic displacement of the link in rad., u is the control input in N m, and d is the input disturbance in N m. Other parameters and their values are given in Table 1. Figure 2 shows the schematic diagram of single-link flexible-joint manipulator. The state-space representation of the system is given by

(89) x ̇ 1 = x 2 , x ̇ 2 = M g L I sin ( x 1 ) K I ( x 1 x 3 ) , x ̇ 3 = x 4 , x ̇ 4 = K J ( x 1 x 3 ) + 1 J ( u + d ) .

Table 1

Parameters of single-link flexile-joint manipulator

Parameters Value Description
M 1 kg Mass of the link
K 1 N m/rad Spring stiffness
L 1 m Length of the link
I 1 kg m 2 Link inertia
J 1 kg m 2 Inertia of the motor
Figure 2 
               Single-link flexible-joint mechanism.
Figure 2

Single-link flexible-joint mechanism.

The state-space equation (Eq. (89)) does not satisfy the matching conditions. Therefore, the coordination of the system (89) should be changed. By considering the state transformation as

[ x ¯ 1 , x ¯ 2 , x ¯ 3 , x ¯ 4 ] T x 1 , x 2 , M g L I sin ( x 1 ) K I ( x 1 x 3 ) , M g L I x 2 cos ( x 1 ) K I ( x 2 x 3 ) T ,

the system will be in the normal form as follows:

(90) x ¯ ̇ 1 = x ¯ 2 , x ¯ ̇ 2 = x ¯ 3 , x ¯ ̇ 3 = x ¯ 4 , x ¯ ̇ 4 = M g L I cos ( x ¯ 1 ) + K I + K J x ¯ 3 + M g L I x ¯ 2 2 K J sin ( x ¯ 1 ) + K I J ( u + d ) .

The fuzzy membership functions are shown in Figure 3. Among all membership functions (triangular, z-shape, trapezoidal, s-shape, sigmoid, and Gaussian), triangular membership functions are selected due to its simplicity, i.e., calculations and modifications of parameters with triangular membership are easy with less computational time. The T–S fuzzy model is obtained by linearizing the system at the center of the membership function of x ¯ 1 and x ¯ 2 as follows:

If x ¯ 1 is about ( 0 ) and x ¯ 2 is about (0), then x ¯ ̇ ( t ) = A 1 x ¯ ( t ) + b 1 u ( t ) + b d d ( t )

If x ¯ 1 is about (0) and x ¯ 2 is about ( π ), then x ¯ ̇ ( t ) = A 2 x ¯ ( t ) + b 2 u ( t ) + b d d ( t ) If x ¯ 1 is about (0) and x ¯ 2 is about ( π ), then x ¯ ̇ ( t ) = A 3 x ¯ ( t ) + b 3 u ( t ) + b d d ( t )

If x ¯ 1 is about ( π ) and x ¯ 2 is about ( 0 ), then x ¯ ̇ ( t ) = A 4 x ¯ ( t ) + b 4 u ( t ) + b d d ( t )

If x ¯ 1 is about ( π ) and x ¯ 2 is about ( π ), then x ¯ ̇ ( t ) = A 5 x ¯ ( t ) + b 5 u ( t ) + b d d ( t ) If x ¯ 1 is about ( π ) and x ¯ 2 is about ( π ), then x ¯ ̇ ( t ) = A 6 x ¯ ( t ) + b 6 u ( t ) + b d d ( t ) If x ¯ 1 is about ( π ) and x ¯ 2 is about ( 0 ), then x ¯ ̇ ( t ) = A 7 x ¯ ( t ) + b 7 u ( t ) + b d d ( t ) If x ¯ 1 is about ( π ) and x ¯ 2 is about ( π ), then x ¯ ̇ ( t ) = A 8 x ¯ ( t ) + b 8 u ( t ) + b d d ( t ) If x ¯ 1 is about ( π ) and x ¯ 2 is about ( π ), then x ¯ ̇ ( t ) = A 9 x ¯ ( t ) + b 9 u ( t ) + b d d ( t )

where

Figure 3 
               Membership function of 
                     
                        
                        
                           
                              
                                 
                                    x
                                    ¯
                                 
                              
                              
                                 1
                              
                           
                        
                        {\bar{x}}_{1}
                     
                   and 
                     
                        
                        
                           
                              
                                 
                                    x
                                    ¯
                                 
                              
                              
                                 2
                              
                           
                        
                        {\bar{x}}_{2}
                     
                  .
Figure 3

Membership function of x ¯ 1 and x ¯ 2 .

A 1 = 0 1 0 0 0 0 1 0 0 0 0 1 9.8 0 11.8 0 , A 2 = A 3 = 0 1 0 0 0 0 1 0 0 0 0 1 86.92 0 11.8 0 , A 4 = A 7 = 0 1 0 0 0 0 1 0 0 0 0 1 9.8 0 7.8 0 A 5 = A 6 = A 7 = A 8 = 0 1 0 0 0 0 1 0 0 0 0 1 86.92 0 7.8 0 , b d = 0 0 0 1 T , b i = 0 0 0 1 T i = 1 , .. . , 9 .

The external disturbance defined as d ( t ) 0.6 0.6 e ( .5 t ) is applied to the system in t = 10 s. The reference signal is determined as x d 0.6 sin ( t ) . The boundaries of the constraint are considered as k c = 0.7 and k x = 0.65 . The design parameters are selected as

κ 1 = 14.2 50 23.2 10 1 , κ 2 = κ 3 = 110.92 50 23.2 10.01 1 ,

κ 4 = κ 7 = 33.8 50 42.8 10 1 , κ 5 = κ 6 = κ 8 = κ 9 = 69.92 50 42.8 10.04 1 .

The adaptation laws are considered as

(91) l ̇ 1 = β 1 2 1 + β 1 2 e 1 2 e 1 2 ( l 1 2 l 2 l 1 ) + e 1 x ˆ 2 l 1 β 1 2 ( k b 2 e 1 2 ) l 1 k = 2 3 x ˆ k 2 l ̇ i = β i 2 1 + β i 2 e 1 2 e 1 2 ( l 1 l i l i + 1 l i ) l i k = 2 i + 2 x ˆ k 2 i = 2 , .. . , 5 .

The stability conditions of the estimation error are indicated by

(92) c 1 l 1 l 1 β 1 2 + 1 + 1 k b 2 e 1 2 > 0 c 2 l 1 + l 1 2 > 0 c 3 l 3 ( l 3 + a 1 ) > 0 c 4 l 3 2 a 4 > 0 c 5 l 4 2 + ( a 1 l 4 ) i = 1 4 a i l i a 4 a 1 2 a 1 ( l 1 l 4 l ̇ 4 l 5 ) + a 1 a 4 l 4 > 0 c 6 l 4 2 a j 1 a j a j 2 a 4 > 0 j = 3 , .. . , 5 c 7 l 4 2 + l 4 a 2 a 1 a 2 a 4 a 2 2 > 0 c 8 l 5 2 l 1 2 l 5 2 l ̇ 5 2 > 1 2 c 9 l 5 2 1 2 > 0 c 10 λ min ( M 5 ) > D 1 .

where

M 5 1 2 β 5 2 ( l 5 2 l 1 2 l 5 2 l ̇ 5 2 ) 0 0 0 0 β 5 2 l 5 2 1 2 β 5 2 l 5 2 0 0 0 0 0 β 5 2 l 5 2 0 0 0 0 0 β 5 2 l 5 2 0 l 5 0 0 0 β 5 2 l 5 2 1 2 .

To find appropriate values for the design parameters and satisfaction of conditions in (92), genetic algorithm (GA) is employed here. Among all five selection techniques (tournament selection, roulette wheel selection, rank selection, uniform selection, and deterministic selection) roulette wheel selection is selected here [56]. The parameters of GA are given in Table 2. The fitness function f ( γ ) is considered as

(93) f ( γ ) γ 1 + γ 3 + γ 4 + γ 5 ,

where γ i β i 2 are the scaling parameters. The design parameters γ i are selected as γ 1 = 1 , γ 3 = 0.9 , γ 4 = 0.81 , γ 5 = 0.93 .

Table 2

Parameters of GA

Parameters Value
Population size 90
Number of generations 100
Crossover rate 0.8
Mutation rate 0.01

The simulation results are illustrated in Figures 48. Figure 4 shows that good trajectory tracking of the proposed method, the NESO in the study of Tran et al. [28], and the ESO in the study of Rsetam et al. [54] is obtained. Moreover, due to the employment of the BLF method in the proposed controller and observer, the output constraint ( k c = 0.7 ) is not violated. Figure 5 compares the tracking errors of the proposed method, the NESO in the study of Tran et al. [28], and the ESO in the study of Rsetam et al. [54]. As shown in the figure, the tracking errors of the proposed method are considerably less than that of the studies of Tran et al. [28] and Rsetam et al. [54]. The maximum amplitudes of the tracking error for the AFESO in the studies of Tran et al. [28] and Rsetam et al. [54] are 0.011, 0.02, and 0.074 rad, respectively. Figure 6 shows that the proposed AFESO estimates states x i ( i = 2 , .. . , 5 ) better than those in the studies of Tran et al. [28] and Rsetam et al. [54]. As shown in Figure 7, the control effort of the proposed method is less than those in the studies of Tran et al. [28] and Rsetam et al. [54]. The adaptive observer gains are shown in Figure 8. Observer gains are bounded and converge to the constant value when the disturbance effect is attenuated.

Figure 4 
               Output trajectory of the proposed method: the NESO in the study of Tran et al. [28], and the ESO in the study of Rsetam et al. [54].
Figure 4

Output trajectory of the proposed method: the NESO in the study of Tran et al. [28], and the ESO in the study of Rsetam et al. [54].

Figure 5 
               Tracking error of different methods.
Figure 5

Tracking error of different methods.

Figure 6 
               States estimation of the proposed method: the NESO in the study of Tran et al. [28], and the ESO in the study of Rsetam et al. [54].
Figure 6

States estimation of the proposed method: the NESO in the study of Tran et al. [28], and the ESO in the study of Rsetam et al. [54].

Figure 7 
               Control inputs of the proposed method: the NESO in the study of Tran et al. [28] and the ESO in the study of Rsetam et al. [54].
Figure 7

Control inputs of the proposed method: the NESO in the study of Tran et al. [28] and the ESO in the study of Rsetam et al. [54].

Figure 8 
               Adaptive observer gains of the proposed method.
Figure 8

Adaptive observer gains of the proposed method.

To compare quantitatively the estimation performance of three methods, the root mean square error (RMSE) is defined as follows:

(94) RMSE = 1 n i = 1 n ( x k ( i ) x ˆ k ( i ) ) 2 k = 1 , , 5 ,

where n is the number of the data samples and x k and x ˆ k are the kth states and its estimation, respectively. The RMSE values of the estimation errors are given in Table 3. As indicated in the table, the RMSEs of the proposed observer are significantly less than those of the studies of Tran et al. [28] and Rsetam et al. [54].

Table 3

RMSE of AFESO, NESO in the study of Tran et al. [28], and ESO in the study of Rsetam et al. [54]

Method x 1 x 2 x 3 x 4 x 5
AFESO 0.0127 0.0553 0.0259 0.016 0.1073
Tran et al. [28] 0.0664 0.162 0.1346 0.0772 0.1114
Restam et al. [54] 0.0258 0.044 0.0881 0.0609 0.1216

In order to evaluate the efficiency of the proposed method in the presence of external disturbance and measurement noise, a white noise with a range of ± 0.3 rad is applied to the system output (Figure 9). Figure 10 illustrates that due to the use of a low-pass filter and an adaptive observer (robustness of the adaptive methods in the presence of noise), the proposed method is able to lessen the effect of the noise. From Figure 11, it can be seen that the proposed method estimates both the states and the disturbances in the presence of measurement noise more accurately than the methods in the studies of Tran et al. [28] and Rsetam et al. [54], where some steady-state errors exist. More importantly, the output constraint is not violated in the proposed method, while the studies of Tran et al. [28] and Rsetam et al. [54] have violated the constraints in some instances.

Figure 9 
               Measurement noise at the output.
Figure 9

Measurement noise at the output.

Figure 10 
               Output trajectory in the presence of external disturbance and measurement noise.
Figure 10

Output trajectory in the presence of external disturbance and measurement noise.

Figure 11 
               States estimation in the presence of external disturbance and measurement noise.
Figure 11

States estimation in the presence of external disturbance and measurement noise.

It should be mentioned that in the studies of Tran et al. [28] and Rsetam et al. [54], the gains are adjusted using the observer bandwidth, which creates susceptibility against measurement noises. On the other hand, in the proposed method, to overcome this problem, the observer gains are adjusted using the proposed adaptation laws. As indicated in Figure 12, the fluctuations in the control efforts of the proposed method are much less than those in the studies of Tran et al. [28] and Rsetam et al. [54] due to the use of a low-pass filter and an adaptive observer.

Figure 12 
               Control input in the presence of external disturbance and measurement noise.
Figure 12

Control input in the presence of external disturbance and measurement noise.

The simulations are carried out using MATLAB® Software on a computer with Intel® Core i7, CPU 2.60 GHz, 8GB DDR3 RAM, and Windows 10 (64-bit). The computation time is 0.163 s for the proposed method, while it is 0.048 s and 0.07 for the studies of Tran et al. [28] and Rsetam et al. [54], respectively. Therefore, all methods can be applied in real-time using conventional computers. Even though the computation time of the proposed method is longer than those in the studies of Tran et al. [28] and of Rsetam et al. [54], it has better estimation accuracy and tracking performance. Moreover, it offers more flexibility in the observer gain design.

7 Conclusion

In this study, an adaptive fuzzy observer controller was designed to control and estimate the states and external disturbances simultaneously in the presence of an output constraint. The adaptive observer gains improved performance of the convectional ESO significantly. One of the advantages of the proposed observer was that it could be applied to higher-order nonlinear systems. The controller was designed based on the command-filtered backstepping scheme and the BLF method to avoid the computational complexity problem of the standard backstepping scheme and to satisfy output constraints. The efficiency of the proposed method in the presence of time-varying external disturbances and output constraints was investigated. The single-link flexible-joint manipulator was considered to demonstrate better performance of the proposed method as compared with the recently published methods in the related literature.

Acknowledgments

The authors thank the Editor-in-Chief and all those in charge of this journal.

  1. Funding information: The authors declare that no funds, grants, or other support were received during the preparation of this study.

  2. Author contributions: Mahtab Delpasand: conceptualization, methodology, software, data curation, validation, formal analysis, investigation, writing – original draft and Mohammad Farrokhi: conceptualization, methodology, validation, formal analysis, writing – review and editing, and supervision.

  3. Conflict of interest: The authors declare that they have no known competing financial interests or personal relationships that could have appeared to influence the work reported in this article.

  4. Ethical responsibilities: The authors confirm that this work is original and has not been published elsewhere, nor is it currently under consideration for publication elsewhere. We declare that the submitted article is a personal academic research article and the authors are not submitting it as an official representative or on behalf of the government. We also declare that none of the authors of the article is employed by a government agency that has a primary function other than research and/or education.

  5. Consent to participate: Informed consent was obtained from all individual participants included in the study.

  6. Data availability statement: The authors confirm that all data generated or analyzed during this study are included in this published article.

References

[1] Ahmad S, Ali A. Active disturbance rejection control of DC–DC boost converter: a review with modifications for improved performance. IET Power Electron. 2019;12(8):2095–107.10.1049/iet-pel.2018.5767Search in Google Scholar

[2] Wei Y, Jia S, Liu K. A survey on anti–disturbance control of switched systems with input saturation. Syst Sci Control Eng. 2020;8(1):241–8.10.1080/21642583.2020.1740113Search in Google Scholar

[3] Chen WH, Yang J, Guo L, Li S. Disturbance–observer-based control and related methods: an overview. IEEE Trans Ind Electron. 2015;63(2):1083–95.10.1109/TIE.2015.2478397Search in Google Scholar

[4] Zhang Y, Chen Z, Sun M. Trajectory tracking control for a quadrotor unmanned aerial vehicle based on dynamic surface active disturbance rejection control. Trans Inst Meas Control. 2020;42(12):2198–205.10.1177/0142331220909003Search in Google Scholar

[5] Han T, Li J, Guan ZH, Cai CX, Zhang DX, He DX. Containment control of multi-agent systems via a disturbance observer-based approach. J Frankl Inst. 2019;356(5):2919–33.10.1016/j.jfranklin.2019.01.034Search in Google Scholar

[6] Sui S, Tong S, Chen CP. Finit-time filter decentralized control for nonstrict-feedback nonlinear large-scale systems. IEEE Trans Fuzzy Syst. 2018;26(6):3289–300.10.1109/TFUZZ.2018.2821629Search in Google Scholar

[7] Ren CE. Adaptive fuzzy disturbance observer-based control for nonlinear uncertain systems with general exogenous disturbances. Int J Fuzzy Syst. 2021;23(5):1453–61.10.1007/s40815-020-01032-3Search in Google Scholar

[8] Hua CC, Wang K, Chen JN, You X. Tracking differentiator and extended state observer-based nonsingular fast terminal sliding mode attitude control for a quadrotor. Nonlinear Dyn. 2018;94:343–54.10.1007/s11071-018-4362-3Search in Google Scholar

[9] Feng H, Guo BZ. Active disturbance rejection control: old and new results. Annu Rev Control. 2017;44:238–48.10.1016/j.arcontrol.2017.05.003Search in Google Scholar

[10] Pu Z, Yuan R, Yi J, Tan X. A class of adaptive extended state observer for nonlinear disturbed systems. IEEE Trans Ind Electron. 2015;62(9):5858–69.10.1109/TIE.2015.2448060Search in Google Scholar

[11] Chan L, Naghdy F, Stirling D. An improved extended active observer for adaptive control of a n−DOF robot manipulator. J Intell Robot Syst. 2017;85:679–92.10.1007/s10846-016-0402-8Search in Google Scholar

[12] Zhang Y, Zhang J, Wang L, Su J. Composite disturbance rejection control based on generalized extended state observer. ISA Trans. 2016;63:377–86.10.1016/j.isatra.2016.03.021Search in Google Scholar PubMed

[13] Madoński R, Herman P. Survey on methods of increasing the efficiency of extended state disturbance observers. ISA Trans. 2015;56:18–27.10.1016/j.isatra.2014.11.008Search in Google Scholar PubMed

[14] Yang G, Yao J, Ullah N. Neuroadaptive control of saturated nonlinear systems with disturbance compensation. ISA Trans. 2022;122:49–62.10.1016/j.isatra.2021.04.017Search in Google Scholar PubMed

[15] Xu Z, Qi G, Liu Q, Yao J. ESO-based adaptive full state constraint control of uncertain systems and its application to hydraulic servo systems. Mech Syst Signal Process. 2022;167:108560.10.1016/j.ymssp.2021.108560Search in Google Scholar

[16] Gandhi RV, Adhyaru DM. Hybrid extended state observer-based control for systems with matched and mismatched disturbances. ISA Trans. 2020;106:61–73.10.1016/j.isatra.2020.06.019Search in Google Scholar PubMed

[17] Castillo A, García P, Sanz R, Albertos P. Enhanced extended state observer-based control for systems with mismatched uncertainties and disturbances. ISA Trans. 2018;73:1–10.10.1016/j.isatra.2017.12.005Search in Google Scholar PubMed

[18] Shi J, Chen X, Yau SS. High order linear extended state observer and error analysis of active disturbance rejection control. Asian J Math. 2019;23(4):631–50.10.4310/AJM.2019.v23.n4.a5Search in Google Scholar

[19] Naghdi M, Sadrnia MA. A novel fuzzy extended state observer. ISA Trans. 2020;102:1–11.10.1016/j.isatra.2019.07.018Search in Google Scholar PubMed

[20] Attar M, Majd VJ, Dini N. Analysis and design of a time-varying extended state observer for a class of nonlinear systems with unknown dynamics using spectral Lyapunov function. J Intell Robot Syst. 2019;94:405–21.10.1007/s10846-018-0935-0Search in Google Scholar

[21] Yang Y, Xie Y, Dai C, Long Z. An improved nonlinear extended state observer with adaptive variable gain. Asian J Control. 2023;25(3):2388–97.10.1002/asjc.2959Search in Google Scholar

[22] Qin Q, Gao G, Zhong J, Ye M. Robust control for uncertain hybrid robot with fluid resistance using unmeasured auxiliary variables-based time-varying gain extended state observer. Int J Control Autom Syst. 2022;20(11):3678–88.10.1007/s12555-021-0814-5Search in Google Scholar

[23] Nie ZY, Zhang B, Wang QG, Liu RJ, Luo JL. Adaptive active disturbance rejection control guaranteeing uniformly ultimate boundedness and simplicity. Int J Robust Nonlinear Control. 2020;30(17):7278–94.10.1002/rnc.5177Search in Google Scholar

[24] Xue W, Bai W, Yang S, Song K, Huang Y, Xie H. ADRC with adaptive extended state observer and its application to air–fuel ratio control in gasoline engines. IEEE Trans Ind Electron. 2015;62(9):5847–57.10.1109/TIE.2015.2435004Search in Google Scholar

[25] He W, Dong Y. Adaptive fuzzy neural network control for a constrained robot using impedance learning. IEEE Trans Neural Net Learn Syst. 2017;29(4):1174–86.10.1109/TNNLS.2017.2665581Search in Google Scholar PubMed

[26] Liao L, Li B, Wang Y, Xi Y, Zhang D, Gao L. Adaptive fuzzy robust control of a bionic mechanical leg with a high gain observer. IEEE Access. 2021;9:134037–51.10.1109/ACCESS.2021.3091132Search in Google Scholar

[27] Pyrkin A, Bobtsov A, Ortega R, Isidori A. An adaptive observer for uncertain linear time-varying systems with unknown additive perturbations. Automatica. 2023;147:110677.10.1016/j.automatica.2022.110677Search in Google Scholar

[28] Tran DT, Jin M, Ahn KK. Nonlinear extended state observer based on output feedback control for a manipulator with time-varying output constraints and external disturbance. IEEE Access. 2019;7:156860–70.10.1109/ACCESS.2019.2949594Search in Google Scholar

[29] Tran DT, Dao HV, Dinh TQ, Ahn KK. Output feedback control via linear extended state observer for an uncertain manipulator with output constraints and input dead-zone. Electron. 2020;9(9):1355.10.3390/electronics9091355Search in Google Scholar

[30] Pajak I. Real-time trajectory generation methods for cooperating mobile manipulators subject to state and control constraints. J Intell Robot Syst. 2019;93(3–4):649–68.10.1007/s10846-018-0878-5Search in Google Scholar

[31] Xu Z, Li L, Yao J, Hu X, Liu Q, Xie N. State constraint control for uncertain nonlinear systems with disturbance compensation. IEEE Access. 2019;7:155251–61.10.1109/ACCESS.2019.2947629Search in Google Scholar

[32] Wei Y, Zheng WX, Xu S. Anti-disturbance control for nonlinear systems subject to input saturation via disturbance observer. Syst Control Lett. 2015;85:61–9.10.1016/j.sysconle.2015.08.006Search in Google Scholar

[33] Yu J, Zhao L, Yu H, Lin C, Dong W. Fuzzy finite-time command filtered control of nonlinear systems with input saturation. IEEE Trans Cybern. 2017;48(8):2378–87.10.1109/TCYB.2017.2738648Search in Google Scholar PubMed

[34] Wang H, Kang S, Feng Z. Finite-time adaptive fuzzy command filtered backstepping control for a class of nonlinear systems. Int J Fuzzy Syst. 2019;21:2575–87.10.1007/s40815-019-00749-0Search in Google Scholar

[35] Yadav VK, Das S. Combination synchronization of fractional order n-chaotic systems using active backstepping design. Nonlinear Eng. 2019;8(1):597–608.10.1515/nleng-2017-0073Search in Google Scholar

[36] Wang X, Yu J. Command filtering and barrier Lyapunov function-based adaptive control for PMSMs with core losses and all-state restrictions. Complexity. 2021;2021:1–12.10.1155/2021/6673568Search in Google Scholar

[37] Wang B, Iwasaki M, Yu J. Command filtered adaptive backstepping control for dual-motor servo systems with torque disturbance and uncertainties. IEEE Trans Ind Electron. 2021;69(2):1773–81.10.1109/TIE.2021.3059540Search in Google Scholar

[38] Cui M, Liu H, Liu W. Extended state observer-based adaptive control for a class of nonlinear system with uncertainties. Control Intell Syst. 2017;45(3):132–41.10.2316/Journal.201.2017.3.201-2770Search in Google Scholar

[39] Zhu Y, Liu J, Yu J, Wang QG. Command filtering-based adaptive fuzzy control of flexible-joint robots with time-varying full-state constraints. IEEE Trans Circuits Syst II Express Br. 2023. 10.1109/TCSII.2023.3295840 Search in Google Scholar

[40] Zhao ZL, Ma P, Chen S. A new nonlinear extended state observer design for output tracking of uncertain nonlinear systems. Adv Control Appl Eng Ind Syst. 2021;3(2):e46.10.1002/adc2.46Search in Google Scholar

[41] Yang Y, Tan J, Yue D. Prescribed performance tracking control of a class of uncertain pure-feedback nonlinear systems with input saturation. IEEE Trans Syst Man Cybern Syst. 2018;50(5):1733–45.10.1109/TSMC.2017.2784451Search in Google Scholar

[42] Lai G, Zhang Y, Liu Z, Chen CP. Indirect adaptive fuzzy control design with guaranteed tracking error performance for uncertain canonical nonlinear systems. IEEE Trans Fuzzy Syst. 2018;27(6):1139–50.10.1109/TFUZZ.2018.2870574Search in Google Scholar

[43] Yao W, Hai Tao Y, Rong G, Dong Yang L, Ningjun F, Zheng X. Fuzzy adaptive sliding mode control of PMSM based on extended state observer. Int J Appl Electromagn Mech. 2020;63(3):391–407.10.3233/JAE-190098Search in Google Scholar

[44] Akraminia M, Tatari M, Fard M, Jazar RN. Designing active vehicle suspension system using critic-based control strategy. Nonlinear Eng. 2015;4(3):141–54.10.1515/nleng-2015-0004Search in Google Scholar

[45] Wu Z, Shi G, Li D, Liu Y, Chen Y. Active disturbance rejection control design for high-order integral systems. ISA Trans. 2022;125:560–70.10.1016/j.isatra.2021.06.038Search in Google Scholar PubMed

[46] Herbst G. Transfer function analysis and implementation of active disturbance rejection control. Control Theory Technol. 2021;19:19–34.10.1007/s11768-021-00031-5Search in Google Scholar

[47] Li S, Yang J, Chen WH, Chen X. Generalized extended state observer based control for systems with mismatched uncertainties. IEEE Trans Ind Electron. 2011;59(12):4792–802.10.1109/TIE.2011.2182011Search in Google Scholar

[48] Sun H, Madonski R, Li S, Zhang Y, Xue W. Composite control design for systems with uncertainties and noise using combined extended state observer and Kalman filter. IEEE Trans Ind Electron. 2022;69(4):4119–28.10.1109/TIE.2021.3075838Search in Google Scholar

[49] Ye H, Song Y. Backstepping design embedded with time-varying command filters. IEEE Trans Circuits Syst II Express Br. 2022;69(6):2832–6.10.1109/TCSII.2022.3144593Search in Google Scholar

[50] Lendek Z, Guerra TM, Babuška R, De Schutter B. Stability Analysis and Nonlinear Observer Design Using Takagi-Sugeno Fuzzy Models. Berlin Heidelberg, Germany: Springer; 2011.10.1007/978-3-642-16776-8Search in Google Scholar

[51] Boyd S, El Ghaoui L, Feron E, Balakrishnan V. Linear matrix inequalities in system and control theory. Philadelphia (PA), USA: Society for Industrial and Applied Mathematics; 1994.10.1137/1.9781611970777Search in Google Scholar

[52] Zhang X, Zong Q, Dou L, Tian B, Liu W. Improved finite-time command filtered backstepping fault-tolerant control for flexible hypersonic vehicle. J Frankl Inst. 2020;357(13):8543–65.10.1016/j.jfranklin.2020.05.017Search in Google Scholar

[53] Ren B, Ge SS, Tee KP, Lee TH. Adaptive neural control for output feedback nonlinear systems using a barrier Lyapunov function. IEEE Trans Neural Net. 2010;21(8):1339–45.10.1109/TNN.2010.2047115Search in Google Scholar PubMed

[54] Rsetam K, Cao Z, Man Z. Cascaded-extended-state-observer-based sliding-mode control for underactuated flexible joint robot. IEEE Trans Ind Electron. 2020;67(12):10822–32.10.1109/TIE.2019.2958283Search in Google Scholar

[55] Khanesar MA, Kaynak O, Teshnehlab M. Direct model reference Takagi–Sugeno fuzzy control of SISO nonlinear systems. IEEE Trans Fuzzy Syst. 2011;19(5):914–24.10.1109/TFUZZ.2011.2150757Search in Google Scholar

[56] Sivanandam SN, Deepa SN. Genetic algorithm optimization problems. Introduction to Genetic Algorithms. Berlin Heidelberg, Germany: Springer; 2008.Search in Google Scholar

Received: 2023-05-16
Revised: 2023-09-16
Accepted: 2023-10-09
Published Online: 2023-11-28

© 2023 the author(s), published by De Gruyter

This work is licensed under the Creative Commons Attribution 4.0 International License.

Articles in the same Issue

  1. Research Articles
  2. The regularization of spectral methods for hyperbolic Volterra integrodifferential equations with fractional power elliptic operator
  3. Analytical and numerical study for the generalized q-deformed sinh-Gordon equation
  4. Dynamics and attitude control of space-based synthetic aperture radar
  5. A new optimal multistep optimal homotopy asymptotic method to solve nonlinear system of two biological species
  6. Dynamical aspects of transient electro-osmotic flow of Burgers' fluid with zeta potential in cylindrical tube
  7. Self-optimization examination system based on improved particle swarm optimization
  8. Overlapping grid SQLM for third-grade modified nanofluid flow deformed by porous stretchable/shrinkable Riga plate
  9. Research on indoor localization algorithm based on time unsynchronization
  10. Performance evaluation and optimization of fixture adapter for oil drilling top drives
  11. Nonlinear adaptive sliding mode control with application to quadcopters
  12. Numerical simulation of Burgers’ equations via quartic HB-spline DQM
  13. Bond performance between recycled concrete and steel bar after high temperature
  14. Deformable Laplace transform and its applications
  15. A comparative study for the numerical approximation of 1D and 2D hyperbolic telegraph equations with UAT and UAH tension B-spline DQM
  16. Numerical approximations of CNLS equations via UAH tension B-spline DQM
  17. Nonlinear numerical simulation of bond performance between recycled concrete and corroded steel bars
  18. An iterative approach using Sawi transform for fractional telegraph equation in diversified dimensions
  19. Investigation of magnetized convection for second-grade nanofluids via Prabhakar differentiation
  20. Influence of the blade size on the dynamic characteristic damage identification of wind turbine blades
  21. Cilia and electroosmosis induced double diffusive transport of hybrid nanofluids through microchannel and entropy analysis
  22. Semi-analytical approximation of time-fractional telegraph equation via natural transform in Caputo derivative
  23. Analytical solutions of fractional couple stress fluid flow for an engineering problem
  24. Simulations of fractional time-derivative against proportional time-delay for solving and investigating the generalized perturbed-KdV equation
  25. Pricing weather derivatives in an uncertain environment
  26. Variational principles for a double Rayleigh beam system undergoing vibrations and connected by a nonlinear Winkler–Pasternak elastic layer
  27. Novel soliton structures of truncated M-fractional (4+1)-dim Fokas wave model
  28. Safety decision analysis of collapse accident based on “accident tree–analytic hierarchy process”
  29. Derivation of septic B-spline function in n-dimensional to solve n-dimensional partial differential equations
  30. Development of a gray box system identification model to estimate the parameters affecting traffic accidents
  31. Homotopy analysis method for discrete quasi-reversibility mollification method of nonhomogeneous backward heat conduction problem
  32. New kink-periodic and convex–concave-periodic solutions to the modified regularized long wave equation by means of modified rational trigonometric–hyperbolic functions
  33. Explicit Chebyshev Petrov–Galerkin scheme for time-fractional fourth-order uniform Euler–Bernoulli pinned–pinned beam equation
  34. NASA DART mission: A preliminary mathematical dynamical model and its nonlinear circuit emulation
  35. Nonlinear dynamic responses of ballasted railway tracks using concrete sleepers incorporated with reinforced fibres and pre-treated crumb rubber
  36. Two-component excitation governance of giant wave clusters with the partially nonlocal nonlinearity
  37. Bifurcation analysis and control of the valve-controlled hydraulic cylinder system
  38. Engineering fault intelligent monitoring system based on Internet of Things and GIS
  39. Traveling wave solutions of the generalized scale-invariant analog of the KdV equation by tanh–coth method
  40. Electric vehicle wireless charging system for the foreign object detection with the inducted coil with magnetic field variation
  41. Dynamical structures of wave front to the fractional generalized equal width-Burgers model via two analytic schemes: Effects of parameters and fractionality
  42. Theoretical and numerical analysis of nonlinear Boussinesq equation under fractal fractional derivative
  43. Research on the artificial control method of the gas nuclei spectrum in the small-scale experimental pool under atmospheric pressure
  44. Mathematical analysis of the transmission dynamics of viral infection with effective control policies via fractional derivative
  45. On duality principles and related convex dual formulations suitable for local and global non-convex variational optimization
  46. Study on the breaking characteristics of glass-like brittle materials
  47. The construction and development of economic education model in universities based on the spatial Durbin model
  48. Homoclinic breather, periodic wave, lump solution, and M-shaped rational solutions for cold bosonic atoms in a zig-zag optical lattice
  49. Fractional insights into Zika virus transmission: Exploring preventive measures from a dynamical perspective
  50. Rapid Communication
  51. Influence of joint flexibility on buckling analysis of free–free beams
  52. Special Issue: Recent trends and emergence of technology in nonlinear engineering and its applications - Part II
  53. Research on optimization of crane fault predictive control system based on data mining
  54. Nonlinear computer image scene and target information extraction based on big data technology
  55. Nonlinear analysis and processing of software development data under Internet of things monitoring system
  56. Nonlinear remote monitoring system of manipulator based on network communication technology
  57. Nonlinear bridge deflection monitoring and prediction system based on network communication
  58. Cross-modal multi-label image classification modeling and recognition based on nonlinear
  59. Application of nonlinear clustering optimization algorithm in web data mining of cloud computing
  60. Optimization of information acquisition security of broadband carrier communication based on linear equation
  61. A review of tiger conservation studies using nonlinear trajectory: A telemetry data approach
  62. Multiwireless sensors for electrical measurement based on nonlinear improved data fusion algorithm
  63. Realization of optimization design of electromechanical integration PLC program system based on 3D model
  64. Research on nonlinear tracking and evaluation of sports 3D vision action
  65. Analysis of bridge vibration response for identification of bridge damage using BP neural network
  66. Numerical analysis of vibration response of elastic tube bundle of heat exchanger based on fluid structure coupling analysis
  67. Establishment of nonlinear network security situational awareness model based on random forest under the background of big data
  68. Research and implementation of non-linear management and monitoring system for classified information network
  69. Study of time-fractional delayed differential equations via new integral transform-based variation iteration technique
  70. Exhaustive study on post effect processing of 3D image based on nonlinear digital watermarking algorithm
  71. A versatile dynamic noise control framework based on computer simulation and modeling
  72. A novel hybrid ensemble convolutional neural network for face recognition by optimizing hyperparameters
  73. Numerical analysis of uneven settlement of highway subgrade based on nonlinear algorithm
  74. Experimental design and data analysis and optimization of mechanical condition diagnosis for transformer sets
  75. Special Issue: Reliable and Robust Fuzzy Logic Control System for Industry 4.0
  76. Framework for identifying network attacks through packet inspection using machine learning
  77. Convolutional neural network for UAV image processing and navigation in tree plantations based on deep learning
  78. Analysis of multimedia technology and mobile learning in English teaching in colleges and universities
  79. A deep learning-based mathematical modeling strategy for classifying musical genres in musical industry
  80. An effective framework to improve the managerial activities in global software development
  81. Simulation of three-dimensional temperature field in high-frequency welding based on nonlinear finite element method
  82. Multi-objective optimization model of transmission error of nonlinear dynamic load of double helical gears
  83. Fault diagnosis of electrical equipment based on virtual simulation technology
  84. Application of fractional-order nonlinear equations in coordinated control of multi-agent systems
  85. Research on railroad locomotive driving safety assistance technology based on electromechanical coupling analysis
  86. Risk assessment of computer network information using a proposed approach: Fuzzy hierarchical reasoning model based on scientific inversion parallel programming
  87. Special Issue: Dynamic Engineering and Control Methods for the Nonlinear Systems - Part I
  88. The application of iterative hard threshold algorithm based on nonlinear optimal compression sensing and electronic information technology in the field of automatic control
  89. Equilibrium stability of dynamic duopoly Cournot game under heterogeneous strategies, asymmetric information, and one-way R&D spillovers
  90. Mathematical prediction model construction of network packet loss rate and nonlinear mapping user experience under the Internet of Things
  91. Target recognition and detection system based on sensor and nonlinear machine vision fusion
  92. Risk analysis of bridge ship collision based on AIS data model and nonlinear finite element
  93. Video face target detection and tracking algorithm based on nonlinear sequence Monte Carlo filtering technique
  94. Adaptive fuzzy extended state observer for a class of nonlinear systems with output constraint
Downloaded on 1.1.2026 from https://www.degruyterbrill.com/document/doi/10.1515/nleng-2022-0344/html
Scroll to top button