Dual band beam steering antenna using branch line coupler network for higher band applications

1 Introduction

Beam-steering antennas are used to mitigate the interferences in multipath environments with good radiation at a certain angle. The uniform distribution of radiated power in a particular direction helps antenna to transmit and receive any information with enhanced throughput in a given channel. Without any mechanical movement based on the phase array principle steering of beam electronically using a switching mechanism is another method for beam-steering [1]. A 2 × 2 array fed microstrip patch antenna with an enhanced gain is proposed by some researchers with a maximum scanning angle of ±17° applicable for 5G communication. In this antenna, author uses a microstrip slot branch circuit of 100 Ω impedance as a power divider at the ground side. At the top side quad rectangular patch is used with connecting arms of 50 Ω impedance [2]. For a millimetre wave (28 GHz) applications few authors have used concept of connected fed array antenna via integrated slot structure. They use single-slot and double-slot connected array antenna with a gain of 9.89 dBi and 12.5 dBi respectively [3], [4]. For satcom applications, an SIW (substrate-integrated waveguide) technique is used by some authors. They use a concept of SGG (substrate-guided ground) which helps antenna to cover a wide scanning angle and good gain. For a massive MIMO antenna, a steerable multiple beam pattern-based antenna as a reconfigurable reflect array is also used [5], [6]. For beam-steering power travels through antenna should be equally divided at output ports. To switch active input ports CMOS-based switches with efficient power are used by some authors for both lower and higher band applications [7]. For adjusting amplitude & phase of beam, antenna on chip array-based configuration with different modes of phase shifter. It allows the expansion of scanning ranges with narrower beams and high data rates [8], [9]. A meta-surface four-layered stack antenna with five different beam steering angles is used as a single unit for satellite communication by some authors [10]. A coupled cascaded transmission line and isolated network leads to transformation of impedance value. Due to this transformation power distribution network provides a narrow beam with good isolation measured at output ports [10], [11]. Generalized joined couplers, and based on Hadamard matric theory concept couplers with switching mechanisms are also designed by different authors for lower band applications [9], [12], [13]. With a compact slot meandering a single antenna gets fed as miniaturized arrangement with eight ports as circularly polarized beam-steering antenna suitable for diversity scheme [14]. A circularly polarized butler matrix based on a Febry-Perot resonator having integration of SIW leads to narrow beam width. Beam-width gets reconfigured after switching to loaded parasitic ring which is applicable for satellite services [15], [16]. Phase reconfigurability is done by calibration of RF input signal, a self-calibration RF circuit has been proposed by some authors to help antenna steer itself at respective angles [17]. After combination of branch line coupler and crossover gets transformed into a compact and reconfigurable beam forming/steering network. For an ultra-wide scanning angle, a 1 × 8 phased array antenna with half-loop topology has been designed by some authors [18], [19], [20], [21]. A beam-steering antenna for radar and mm-wave applications has been designed using a branch line coupler (BLC) and a monopole antenna using Rogers-RT Duroid 5,880 (Ꜫr = 2.2 & dissipation factor of 0.0009). Formation of beam-steering antenna is a combination of BLC network with fed monopole antenna at load. A monopole antenna is designed using DGS technique and its design steps/flow with complete responses is discussed in Section 1. Another Section 2 shows analysis and design of branch line coupler with transition of lower to higher operating bands. Section 3 shows analysis of beam-steering antenna after excitation of both ports one and two simultaneously and independently. In this article some important results of antenna diversity parameters are discussed with beam-steering mechanism at wide scanning angle for respective operating frequencies.

2 Development of monopole antenna

For a beam-steering antenna, development of an elementary antenna is required to fed at respective ports of BLC network. A compact horn structured monopole antenna using DGS having size of 5 × 5 × 0.8 mm³ has been proposed as the fed antenna. To maintain miniaturization, DGS plays a vital role in designing this antenna. Figure 1. Shows the designing steps with the proposed antenna’s structure. Fed line of 50 Ω impedance is used having a width of 1.8 mm and length of 1.6 mm and these parameters are calculated at an operating frequency of 40 GHz. In step 1 a T-shaped patch structure is used with full ground as a microstrip patch antenna having the size of 5 × 5 × 0.8 mm³. It is operating at 52 GHz and beyond but it does not cover the desirable bands. In step 2, T shaped patch gets converted into a horn structure operating at same band with an improved reflection coefficient (|S₁₁|). In next step, a rectangular slot of 0.2 mm is formed before the horn structure and its behaviour seems to be same as horn structure antenna. Then a monopole or partial ground structure is used in step 4 where dual band is achieved where, 1st band operating at 20 GHz and 2nd band at 60 GHz with unsatisfied return loss. In the next step 5, a uniform stair type structure is used as DGS but its response seems to be same as last designed antenna. A half-circular ring of width 0.5 mm gets added to the stair-based structure to enhance the technique of DGS to reach the desirable operating band. Keeping the patch same as in step 3 and enhancing the technique of DGS leads to final proposed antenna. This antenna covers a dual band with cut-in frequency (Fc) of 24.4 GHz and 39.8 GHz with reflection coefficients of −23 dB and −16 dB respectively. B.W % of 42 % and 14.2 % are observed at both respective bands (20.2–31.2 GHz; 36.6–42.2 GHz). The responses of reflection coefficient (|S₁₁) at operating frequencies for all designing steps including the proposed antenna are shown in Figure 2. VSWR (voltage standing wave ratio) versus frequency response is shown in Figure 3a. covers the of VSWR in between one and two for both operating bands. The desirable range of VSWR should be under two considered as good match for any antenna applications. The plot of |S₁₁| gets integrated with VSWR plot at respective bands for better comparison as shown in Figure 3a. Although directivity & antenna gain are closely related, antenna gain also considered as antenna’s efficiency. The ratio of the intensity in a particular direction and the radiation intensity, that would be obtained if the power accepted by antenna were radiated isotopically is termed as gain. Antenna gain ranges from 3.2 to 4 dBi and 4.8–4.2 dBi having a maximum gain of 4.8 dBi with an average radiation efficiency of 95 % for both operating bands. This efficiency shows the relation between the maximum power radiated by an antenna with power fed at excited port. The response of these parameters is shown in Figure 3b. The behaviour of conducting element/radiator depends on radio frequencies and due to dissipation factor of any substrate there is a phase shift in radio signals. Due to this, there is a rate of change in the transmission signal with some phase shift termed as group delay (GD). The response of GD at a respective band with |S₁₁| response is shown in Figure 4a. At maximum return loss, there is a peak in the group delay response for both bands. At 0 dB GD is 40 ps (pico-second), at −24 dB GD 170 ps and at −15 dB GD is 170 ps, which clearly shows the dependent relationship between the group delay and reflection coefficient.

Figure 1:

Design flow of proposed antenna at different designing steps with its parametric values. Proposed antenna (a) monopole antenna prototype (b) DGS structure with A₁ = 3 mm, H = 2 mm, a = 0.5 mm (c) patch structure with L₁ = 1.8 mm, L₂ = 1.6 mm, L₃ = 1.4 mm, L₄ = 1.2 mm, L₅ = 3 mm and g = 0.4 mm.

Figure 2:

Responses of S-parameter [S₁₁] versus frequency of each designing steps and final proposed antenna.

Figure 3:

(a) Response of VSWR with S₁₁ at resonating frequency (b) peak gain and radiation efficiency plot of monopole antenna.

Figure 4:

(a) Response of group delay and S₁₁ for monopole antenna (b) impedance response with S₁₁ at respective operating frequencies.

An input impedance of 50 Ω is used for calculating the dimension of width and length of fed transmission line connected to the conducting element. At this impedance value power gets excited as input and maximum amount of power gets reflected back as a return loss/reflection coefficient. At maximum return loss impedance value should be approximately to 50 Ω for better matching and efficiency of antenna. The observed value of impedance ranges between 40 and 70 Ω at both bands as shown in Figure 4b. Figure 5 shows the radiation pattern at E-plane and H-plane with co and cross (X) polarization. Co-polarized pattern is observed when the transmitting and receiving antenna is in same direction or in same plane while a cross-polarized pattern occurs when both transmitting and receiving antenna is perpendicular to each other.

Figure 5:

Radiation pattern (a) E-plane (phi = 0°) (b) H-plane (phi = 90°) of proposed monopole antenna.

The electric field plane & magnetic field plane should be perpendicular to each other as phi = 0° and phi = 90° respectively. For this monopole antenna the level of cross-polarization is circularly polarized. The maximum amount of a current is distributed over the conducting surface of monopole antenna is also measured as an electric field (V/m) at respective operating frequencies. The maximum amount of electric field at 24 GHz and 40 GHz is 15000.91 V/m and 20569 V/m respectively as shown in Figure 6a and b.

Figure 6:

Electric field plot (a) at 24 GHz (b) 40 GHz.

3 Design of 90° hybrid branch line coupler

Before approaching to beam-steering network, a four-port directional coupler is designed with a size of 12.11 × 13.39 × 0.8 mm³ (0.96λ₀ × 1.07λ₀ × 0.06λ₀). All four ports are matched with Z₀ characteristics impedance and it behaves as a transmission line to establish a connection between each port via Z 0 2 characteristics impedance transmission line horizontally and Z₀ impedance vertically. An input power gets excited from port 1 and it gets divided as half power transmitted to port 2 directly and half power gets transmitted with a phase difference of 90° towards the coupled port 3. No power will travel through port 4 hence it is considered an isolated port. This network is lossless and also called a branch line coupler (BLC). Figure 7a shows a view of BLC with characteristics impedance and port allocation where, Figure 7b shows the parametric view of branch line coupler. The connecting transmission line (horizontally and vertically) between each port is at λ/4 distance to maintain the phase difference of 90° between ports 2 and 3 and due to this it is also known as a quadrature hybrid coupler. Designing BLC at the higher operating frequency is more challenging because length and width of the transmission line at Zo and Zo/1.41 impedance gets reduced and overlapping between connecting arms may occur. So, by maintaining the separation between the transmission line with λ/4 value for good matching, the designing process of BLC should be at a lower operating frequency. As BLC covers a large wide bandwidth and multiple frequencies at 2nd and 3rd harmonics or iterations. Utilizing this concept the parameters of all transmission lines at each impedance value are calculated at 10 GHz and the response of this designed BLC is shown in Figure 8a. At the second harmonic, this BLC is operating at 24 GHz and the response of s-parameters for each port is shown in Figure 8b. At the designed operating frequency (10 GHz) a perfect isolation at port 4 (S₁₄) is observed with a return loss of –18 dB at port 1 (S₁₁). The equal power gets distributed at port 2 (S₁₂) and port 3 (S₁₃) with −3 dB. At higher frequency BLC covers a wide B.W from 22 to 45 GHz having S₁₁ ≤ −10 dB and S₁₂ of −3 dB at 24 GHz. Also analysed the group delay (GD) response dependent to rate of change of transmitted signal at desired operating frequency. At isolated port, GD₁₄ or GD₄₁ achieves maximum delay because RF signal traveling through these ports should take much time to reach. Input signal is excited at port 1 and gets distributed equally over the port 2 and 3 with phase shift of 90° and response of group delay plot seems equal at 10 GHz. Simulated response of GD is shown in Figure 9a for lower operating frequency (10 GHz). For higher frequency (24 GHz) delay at through port is of minimum value and delay at input and coupled ports are almost equal. The response of GD at isolated port is completely different from other ports. The response plot of group delay at higher frequency is shown in Figure 9b.

Figure 7:

Branch line coupler design (a) with impedance parameter (Zo = 50 Ω) (b) with parametric value as L = 12.11 mm, L₁ = 2.34 mm, L₂ = 3.81 mm, L₃ = 4.49 mm, W = 13.39 mm, W₁ = 4.49 mm, W₂ = 4.41 mm and W₃ = 2.34 mm.

Figure 8:

S-parameter response of BLC (a) at 10 GHz [lower operating frequency] (b) at 24 GHz [higher operating frequency].

Figure 9:

Group delay responses at each port for (a) 10 GHz BLC (b) 24 GHz BLC.

The proposed BLC gets transformed into a dual BLC network as shown in Figure 10 having size of 12.11 × 22.29 × 0.8 mm³. This BLC network covers a bandwidth of 22.4–26.7 GHz and 29.8–45 GHz with B.W % of 17.55 % and 40.6 % respectively at each excited port (S₁₁, S₂₂, S₃₃ & S₄₄). The simulated response of S-parameters for all the ports is shown in Figure 10. The reflection coefficient for 1st and 2nd bands is ≤−10 dB respectively.

Figure 10:

Transformed BLC network with reflection coefficient response at each excited port.

4 Beam steering network fed array antenna

The monopole antenna using DGS technique gets fed at port 2 and port 3 of the dual BLC network is termed as a beam-steering antenna. The prototype view of proposed beam-steering antenna with its dimension and port allocation is shown in Figure 11(a) as top and bottom view, similarly the fabricated antenna is also represented in Figure 11(b). For verifying multiple antenna characteristics an analysis has done by exciting each port simultaneously. The return loss at desirable operating frequency with wide bandwidth shows a better match for any antenna. The simulated and measured reflection coefficient plot at operating frequencies is shown in Figure 12(a). The simulated |S₁₁| & |S₂₂| ≤−10 dB for both band by covering B.W of 22.4–26.9 GHz and 29–45 GHz respectively. The measured reflection coefficient at port 1 and port 2 is very close to the simulated value. It is observed that second operating band shows maximum simulated |S₁₁| of −52 dB and measured of −40 dB at port 1 & 2 respectively. Similarly, simulated & measured |S₁₁| of −25 dB and 28 dB for the first band. The simulated isolation or insertion loss in between the adjacent identical fed antenna as |S₁₂| and |S₂₁| is greater than 22 dB for 1st band and 25 dB for another band. The measured isolation is greater than 15 dB and 20 dB respectively for both bands is found. The response plot for the same is shown in Figure 12(b).

Figure 11:

Beam steering antenna as top and bottom view (a) prototype view (b) fabricated/hardware view.

Figure 12:

Simulated and measured response of (a) reflection coefficient (b) isolation between adjacent fed antenna.

The decoupling between each antenna with a good match is observed and envelope correlation coefficient (ECC) is affected by mutual coupling. If the value of ECC is close to zero then coupling will be very low. Using scattering parameter characteristics value of ECC is calculated using equation (1). For two adjacent antenna elements, S_XX is identified as |S₁₁|, S_XY as S₁₂, S_YX as S₂₁, and S_YY as S₂₂ used in equation (1). The simulated ECC is very close to zero and the measured ECC is 0.025. Based on ECC diversity gain (DG) is also measured using equation (2). This antenna also behaves as a MIMO antenna and its diversity parameters get affected in terms of signal-to-interference ratio. Due to dual antenna at

the fed port, the radiation from each element leads to interreferences and it will affect the overall performance of the antenna. Diversity gain shows the amount of power that gets reduced in any diverse environment without decaying the antenna performances. For a better match, DG should be 10 and this antenna shows the simulated value of 9.88 and measured as 9.255. The response of ECC and DG as measured and simulated is shown in Figure 13(a). The mean effective gain (MEG) shows that in multiple path surroundings, the amount of power gets received by an antenna. Basically, it is a ratio of received power to incident power and it is calculated using equation (3). MEG at ports 1 and 2 for two radiating elements is calculated using equations (3a) and (3b) respectively and resultant MEG is found using equation (3c). For better performance of antenna in a multipath fading MEG should be less than −3 dB. For this antenna resultant MEG is reported at 0 dB having MEG₁ & MEG₂ ≤ −5 dB. The response plot of MEG is shown in Figure 13(b).

Figure 13:

(a) Simulated and measured response of ECC and DG (b) response plot of MEG.

After analysis of decoupling characteristics for two adjacent antenna in a multipath environment this antenna is used to steer the beam pattern with phase shift of 90°. The simulated and measured response of reflection coefficient after exciting each port independently (in alternate mode) is shown in Figure 14(a) covering dual bands. The reflection coefficient ≤−10 dB at port 1 and port 2 for both measured and simulated. The maximum return loss is −50 dB at 24 GHz, −42 dB at 33 GHz, and −35 dB at 39 GHz respectively as shown in Figure 14(a) by covering the required bandwidth for Radar and mm-Wave applications with B.W % of 17.40 % and 40 % respectively. Antenna gets tested via Vector Network Analyzer (VNA) and the measured response at excited port is shown in Figure 15. The antenna testing set up in an anechoic chamber is also presented in Figure 15. Measured and simulated response of VSWR is observed with very slight variations in responses for both excited ports as shown in Figure 14(b). For 1st band, the values of VSWR lies in between one and 2 & for 2^nd band it lies in between one and 1.5. Simulated antenna peak gain of 9 & 9.5 dBi for both band 1 and 2 respectively at port 1. The measured peak gain at port 1 is 8.3 and 8.5 dBi. At port two peak gains of seven dBi & 7.2 dBi are observed as simulated values for both bands, while the measured peak gain of 6.8 dBi and 7.5 dBi is found.

Figure 14:

Measured and simulated (a) return loss response (b) VSWR plot.

Figure 15:

Antenna measuring set up with VNA and in anechoic chamber.

Radiation efficiency for both bands is reported in between 83 and 89 % as simulated and 81–86 % as measured. The response plot of peak gain and radiation efficiency is shown in Figure 16(a). Maximum group delay response is shown in nano second (ns) at respective operating frequencies where, maximum return loss is observed. The response of group delay is shown in Figure 16(b). Beam-steering mechanism is analysed by switching each port alternatively at respective resonating frequencies. At 24 GHz the radiation pattern or beam is directed towards the +45° and −75° when port 1 gets excited and if port 2 gets excited the similar beam gets steered by 90° at −45° and +75° respectively. Similarly, at 33 GHz beam pattern gets steered by 90° shifts after exciting each port independently, with scanning angles of ±45° and 180°. At 36 GHz pattern is uniformly directed towards −45° angle and gets steered after switching to port 2 by +45°. Scanning angles at 39 GHz are ±15° and 45° respectively with the same radiation intensity is found. A directional beam gets shifted by 90° at 42 GHz with a scanning angle of ±10°. The simulated and measured beam-steered pattern at the respective frequencies for port 1 and port 2 is shown in Figure 17. Through the conducting surface of the beam-steering antenna distribution of current in V/m is shown in Figure 18 when port 1 is active. The maximum current travels from port 1 and it gets distributed equally to both fed antenna as an array. Port 2 receives a very small amount of current and behaves as an isolated port. The same mechanism will work when port 2 gets excited and port 1 behaves as an inactive or isolated ports. For 24 GHz at port 1 and port 2, maximum current distributed over conducting surface is 13201 V/m. Similarly at 39 GHz maximum current over the conducting surface is 11441 V/m after exciting each port independently. The observed field plot over the surfaces of radiating element is shown in Figure 19. The response of the proposed 2 × 2 fed array BLC network-based beam-steering antenna is summarized in Table 1. A good insertion (|S₂₁| is greater than 20 dB) and return loss (|S₁₁| ≤ −10 dB) is achieved at port 1 and port 2 respectively. It shows average radiation efficiency of 88 % with B.W % of 17 and 40 % for dual-band applicable for Radar and mm-Wave applications. Adequate gain is achieved for both operating bands as 9 dBi and 10.2 dBi respectively. For diversity scenario ECC, DG, and MEG are also checked and acceptable value has been achieved. A comparison between final proposed antenna with different latest research work has been also done. Based on antenna design techniques and their performing parameters Table 2 is formed. The proposed antenna is applicable for multiple applications with good gain and enhanced isolation having wide scanning angles.

Figure 16:

Measured and simulated response at port one and port 2 (a) peak gain & radiation efficiency (b) group delay.

Figure 17:

Beam-steering mechanism response at 24 GHz, 33 GHz, 36 GHz, 39 GHz and 42 GHz respectively.

Figure 18:

Current distribution over the antenna at 24 GHz for excited port 1 & 2 respectively.

Figure 19:

Current distribution over the antenna at 39 GHz for excited port 1 & 2 respectively.

5 Conclusions

A 2 × 2 hybrid coupler based a beam-steering antenna has been proposed in this research article applicable for the Radar and millimetre wave applications. It covers the dual band having the bandwidth of 22.6–26.89 GHz & 30–45 GHz with radiation efficiency of 88 % and gain of 9 dBi and 10.2 dBi respectively. All observed response of the proposed antenna gets summarized in Table 1. At different operating frequencies a directional beam pattern at distinct angles is found with wide scanning angles. Desirable response of insertion loss, ECC, DG and MEG are observed which makes this antenna as a good candidate in multipath environment. A comparison has been made with some latest work done by some authors gets summarized in Table 2. Due to narrow beam pattern at different angles antenna is applicable for higher frequency applications by covering Radar, ISM-III band and millimetre wave applications.