A novel miniaturized microstrip filtering power divider with high selectivity based on composite right/left-handed (CRLH) concept

1 Introduction

Passive microwave devices based on the unusual properties of the metamaterial structures and predominantly composite right/left-handed (CRLH) structures have recently attracted much attention [1], [2], [3], [4], [5], [6], [7], [8], [9]. The CRLH structures are artificial materials with simultaneously negative permittivity and permeability, which are extensively utilized to modify the performance of passive microwave circuits like band-pass filters (BPF), diplexers, couplers, and power dividers (PD) [1], [2], [3], [4], [5], [6], [7], [8], [9]. The CRLH structures could be implemented by resonance or non-resonance configurations [1], [2], [3], [4], [5], [6], [7], [8], [9]. The resonant-type approach is more appropriate for realizing the BPFs and PDs [1], [2], [3], [4], [5], [6], [7], [8], [9]. The split-ring resonators (SRRs) and the complementary split-ring resonators (CSRRs) are two of the most famous elements of this type of structure [1], [2], [3], [4], [5], [6], [7], [8], [9].

Filtering power divider (FPD) is a key microwave passive component with both the power divider function and filtering function, has attracted much attention in the modern wireless communication system. Recently, different types of the FPDs have been introduced based on the several techniques to realize high performances although there are still many challenges in fabricating these devices [10], [11], [12], [13], [14], [15], [16], [17], [18], [19], [20], [21], [22], [23]. These reported FPDs to have disadvantages in terms of considerable size, high-insertion loss, high cost and complex manufacturing. However, there is still an essential requirement to design FPDs with miniaturized size, low fabrication cost, low loss and high selectivity.

In this paper, a novel compact microstrip FPD according to the CRLH concept is introduced. The proposed microstrip FPD is implemented based on the combination of two CRLH unit cell which is grounded by means of via. The designed CRLH unit cell consists of fractal/meander complementary split-ring resonator (FMCSSR) etched in the ground plane and an interdigital capacitor between two transmission lines (TL) on the upper metal layer. By using this configuration, an ultra-compact microstrip power divider with filtering response, low cost, easy fabrication, low loss, high selectivity, and high Q-factor has been obtained. To confirm the design procedure, the introduced microstrip FPD is fabricated and measured. A suitable agreement between the simulated and measured results has been achieved.

2 Design procedure of the proposed microstrip FPD

The configuration and the equivalent circuit model of the proposed CRLH unit cell is shown in Figure 1. The proposed CRLH unit cell consists of an interdigital capacitor that is loaded between two microstrip transmission lines (TL) on the upper metal layer and the fractal/meander complementary split-ring resonator (FMCSRR) on the ground plane (underneath the conductor strips and interdigital capacitor). Several novel techniques have been used to miniaturize conventional complementary split-ring resonator (CSRR) [4], [5]. The fractal and meander techniques are also attractive for reducing the size of the traditional CSRR [6], [7]. Accordingly, a new tape of CSRR structure has been introduced, called the FMCSRR. In the FMCSRR configuration, the conventional transmission line is replaced with a fractal slot line and the meander slot line is also etched inside the inner space of the ring. The FMCSRR structure is applied for the synthesis of ε-negative metamaterial configuration. It has been modeled as a parallel resonant tank with inductance L_r and capacitance C_r. The capacitance C_r is the capacitance of a square-shaped plate surrounded by a metallic plane. The inductance L_r is the inductance of the metallic strip between the ring slots. Consequently, using these techniques, the resonance frequency of the FMCSRR configuration is decreased compared to the traditional CSRR structure with the exact sizes. In other words, by applying the fractal and meander techniques, the physical size of the traditional CSRR configuration could be decreased without varying its central frequency. As previously mentioned, the FMCSRR structure implements the negative permittivity (ε < 0) media in the vicinity of its resonant frequency, obtained by applying an axial time-varying electric field. Therefore, the FMCSRR configuration can be excited by etching in the ground plane of a microstrip transmission line, which causes the signal to be inhibited in a narrow band [6], [7]. An extra component is needed to synthesize the left-handed medium to provide the negative permeability (μ < 0). The required negative effective permeability can be achieved by etching series interdigital capacitors in the host transmission line [6], [7]. Consequently, the double negative (DNG) medium (ε < 0 & μ < 0) consisting of the FMCSRR structure and interdigital capacitor exhibit a band-pass response in the narrow band above FMCSRRs resonance [6], [7]. Accordingly, the proposed microstrip band-pass filter (BPF) designed based on the CRLH concept is shown in Figure 2. The T-circuit model of this microstrip filter is shown in Figure 3(a). The simulated frequency responses of the proposed microstrip filter in comparison with the frequency responses of the circuit model are also illustrated in Figure 3(b), which verify our prediction. The FMCSRR structures in the equivalent-circuit model are modeled by the resonant tank, which described by the capacitance C_r and the inductance L_r. The interdigital capacitors are modeled through the capacitance C_g. The vertical grounded stubs with via holes can also be considered as shunt inductance L_via. L is the inductance of the line, whereas C_TL is the line capacitance and the fringing capacitance of the gaps, which models the electric coupling between the line and the FMCSRR structures. In practice, because of the small width of the transmission line, the capacitor of the TL is negligible and not affect the characteristics of the transmission zeros unless a small shift of the zeros within the stopbands. Accordingly, for simplicity in calculation, the effect of the C_TL is neglected. Also, for simplicity, the effects of the parasitic shunt capacitance and series inductance of the interdigital capacitor can be neglected. The element values of the circuit model for the designed microstrip filter are: the series capacitance of the interdigital capacitor is approximately C_g = 0.6 pF and the shunt inductance of the via hole is approximately L_via = 0.15 nH. The capacitance and inductance of the TL are C_TL = 0.01 pF and L_TL = 6.4 nH, respectively. Also element values of the FCSRR are C_r = 2 pF and L_r = 0.52 nH.

Figure 1:

Configuration of the proposed CRLH unit cell (a) top layer, (b) bottom layer and, (c) equivalent-circuit model.

Figure 2:

Configuration of the proposed equal microstrip band-pass filter (BPF) (a) top layer and, (b) bottom layer.

Figure 3:

The circuit model of the proposed microstrip filter and the simulated frequency responses in comparison with the frequency responses of the circuit model. (a) Equivalent-circuit model and (b) the frequency responses of the circuit model and simulated of the proposed microstrip BPF.

The extracted real part of the permittivity (ε) and permeability (μ) for the proposed microstrip filter is shown in Figure 4. As confirmed in this figure, a negative permittivity and a negative permeability has been occurred in the vicinity of f_r = 2.42 GHz, which are the resonance frequency of the FMCSRR structure. This figure verifies that by combining an interdigital capacitor which is loaded between two microstrip transmission lines (TL) on the upper metal layer and the FMCSRR on the ground plane, an ε-negative material and μ-negative material in the vicinity of the resonance frequency of the FMCSRR structure are achieved. This means that, a double negative medium can be achieved by using the proposed CRLH structure.

Figure 4:

Extracted (a) real part and (b) imaginary part of the permittivity (ε) and permeability (μ) for the introduced band-pass filter.

Figure 5 shows the configuration of the suggested microstrip FPD. The proposed design is implemented by cascading two CRLH unit cells and a grounded via hole. The grounded via hole has been used to improve frequency selectivity and out-of-band rejection of the proposed microstrip FPD. The dimensions of the introduced microstrip FPD are listed as follows: w₁ = 1.6 mm, w₂ = 2.4 mm, w₃ = 1.13 mm, w₄ = 1.13 mm, w₅ = 5.2 mm, w₆ = 1.2 mm, l₁ = 6.4 mm, l₂ = 1 mm, l₃ = 1.6 mm, l₄ = 1.6 mm, l₅ = 5.2 mm, l₆ = 1.2 mm, l₇ = 1.4 mm, S = 0.2 mm, c = 0.2 mm, d = 0.8 mm. The simulated frequency responses of the equal microstrip FPD have been displayed in Figure 6. The most important feature of the proposed structure is its small size. Another advantage of the proposed microstrip FPD is the ease of frequencies shifting. In Figure 7, the simulated frequency responses of the proposed microstrip FPD by changing the dimension of the FMCSRR structure are shown. As demonstrated in this figure, by resizing the FMCSRR structure, the center frequencies of the passband can be simply adjusted and the proposed FPD can be worked in the desired locations. By resizing the length and width of the interdigital capacitor and changing the number of the fingers in the interdigital capacitor, the return loss has been modified and the center frequency can be tuned. By varying the distance between two FMCSRR structure, the bandwidth could be adjusted. Furthermore, the bandwidth and the return loss can be easily regulated by resizing the w₁. As well as, the phase difference between the two output ports over the entire operating bandwidth has been depicted in Figure 8. The simulated phase difference between the two output ports in the passband covering 2.38–2.45 GHz for the suggested microstrip FPD is within −4° to −2.8°.

Figure 5:

Configuration of the proposed equal microstrip FPD (a) top layer and, (b) bottom layer.

Figure 6:

Simulated results of the proposed equal microstrip FPD.

Figure 7:

Simulated frequency responses of the proposed equal microstrip FPD by resizing the FMCSRR structure.

Figure 8:

Simulated differential phase of the proposed equal microstrip FPD.

3 Fabrication and measurement

According to the principle stated in the previous section, the proposed equal microstrip FPD has been fabricated and tested. Figure 9 shows the photographs of the fabricated microstrip FPD with equal power division ratios. The introduced equal microstrip FPD was implemented using Rogers RO4003C substrate with a dielectric constant of 3.55, loss tangent of 0.0027, and thickness of 0.508 mm. A 50 Ω microstrip line is used on both sides for the impedance matching and for measurement, which the width of these microstrip feed lines has been calculated using the ADS line calc. The proposed configuration has been simulated using the ADS electromagnetic simulator, and experimental measurements have been attained by utilizing a vector network analyzer (VNA) Rohde & Schwarz, zvk. A good matching between the measured and simulated results has been achieved. The center frequency of the manufactured microstrip FPD is 2.42 GHz which is appropriate for WLAN applications. The total size of the presented equal microstrip FPD is about 11.6 mm × 5.2 mm (0.17 λ_g × 0.07 λ_g). Figure 10 shows the simulation and measurement results of the designed microstrip FPD. The insertion loss of the fabricated FPD is around 1.4 dB, and the return loss is better than 18 dB. The 3 dB fractional bandwidth (FBW) is approximately 2.9 % from 2.38 to 2.45 GHz. A comparison between the proposed microstrip FPD and other FPDs is given in Table 1. As can be seen in Table 1, it is clear that the proposed microstrip FPD has the advantages of compact size, low insertion loss, high Q-factor, high selectivity, easy band-pass frequency shifting, and easy fabrication.

Figure 9:

Photograph of the fabricated equal microstrip FPD (a) top layer, and (b) bottom layer.

Figure 10:

Measured and simulated results of the fabricated equal microstrip FPD.

4 Conclusions

This paper presents a new topology for the compact microstrip filtering power divider (FPD) design with high selectivity and band-pass filtering response. The design procedure is based on the composite right/left-handed (CRLH) concept. The proposed configuration is implemented by cascading two CRLH unit cells and a grounded via hole. The proposed CRLH unit cell is realized by an interdigital capacitor loaded between two transmission lines (TL) on the upper metal layer and the fractal/meander complementary split-ring resonator (FMCSRR) on the ground plane. The grounded via hole has been used to improve frequency selectivity and out-of-band rejection of the proposed microstrip FPD. The introduced microstrip FPD has been fabricated and tested. Good agreement between the simulated and measured results is achieved. The whole dimension of the proposed microstrip FPD is only about 0.17 λ_g × 0.07 λ_g, which confirms that the introduced microstrip FPD has a miniaturized dimension. In addition to compact size, the proposed microstrip FPD has many advantages in terms of low insertion loss, high return loss, low fabrication cost, high Q-factor and high selectivity.