A metamaterial loaded hybrid fractal multiband antenna for wireless applications with frequency band reconfigurability characteristics

1 Introduction

2 Antenna design and configuration

This part of paper mentioned the evolution stages of proposed slotted multiband antenna (loaded with metamaterial cell and fused fractal geometry) and respective S-parameter (simulated S₁₁) are illustrated in Figures 1 and 2 respectively. Proposed design is constructed on FR4 dielectric substrate (ε_r = 4.4, tanδ = 0.02) with compact dimensions of 29.5 × 22 × 1.6 mm³.

Figure 1:

Design evolution stages of proposed design.

Figure 2:

Simulated S₁₁ of proposed antenna for each design stage.

The multiband and frequency band reconfigurability characteristics in the proposed structure are achieved by using following design configuration stages, design stage-I: during design evolution stage-I, antenna consist an octagonal shape radiating section fed with trapezoidal shape microstrip line covering UWB single resonant mode with impedance bandwidth 10.33 GHz (3.12–13.45 GHz, 124.68%) in simulation mode as illustrated in Figure 1 (stage-I) and Figure 2 respectively.

Design stage-II: In this stage inserting the octagonal shape slot with three duplication sections (six segments) of fused hybrid fractal (first iteration Moore and Koch curve) geometry inside the radiating patch to make the antenna design to resonate at triple band state with operating bandwidth 2.68 GHz (3.94–6.62 GHz, 50.75%), 1.75 GHz (8.22–9.97 GHz, 19.24%) and 0.62 GHz (13.02–14.14 GHz, 8.25%) for C band WLAN, Lower X band and Lower Ku band wireless standards respectively, as represented in Figure 1 (stage-II) and Figure 2. The slotted hybrid fractal formation in radiating part is responsible to create the current perturbation effect in proposed design that drives the multiband (triple) band characteristics.

Design stage-III: Under this design evolution stage, loading the two metamaterial rectangular SRR cells within radiating patch to achieve an additional resonant bands at wireless standards Lower C, WLAN, Lower X and Lower Ku band with impedance bandwidth 0.91 GHz (3.76–4.67 GHz, 21.59%), 1.39 GHz (5.12–6.51 GHz, 23.90%), 2.24 GHz (7.71–9.95 GHz, 25.37%) and 2.56 GHz (12.89–15.45 GHz, 18.07%) respectively (during simulation process), as depicted in Figure 1 (stage-III) and Figure 2. It is observed that the proposed SRR cells are liable to create an additional narrow band (as compared to triple resonant bands in Design stage-II) at Lower C band wireless communication mode because of negative permeability characteristics of metamaterial SRR. The solenoidal conducting current is passing through the rectangular SRR rings and obtain the magnetic response regarding additional resonant band.

Design stage-IV: By placing the seventh section of hybrid fractal geometry in radiating part to obtain the hepta band configuration at resonant bands 3.5, 4.41, 5.4/5.8, 8.26, 10.48, 13.35 and 14.42 GHz for wireless applications WiMAX (2.86–3.81 GHz), Lower C band (4.25–4.76 GHz), WLAN (5.01–7.62 GHz), Lower X-band (7.75–10.15 GHz), Upper X band (10.43–11.16 GHz), Lower Ku band (12.84–13.98 GHz) and Middle Ku band (14.12–15.89 GHz) respectively, as indicated in Figure 1 (stage-IV) and Figure 2. The respective seventh fused fractal structure is adjoined with feedline by using conducting strip line with RF switching element PIN diode inside it (Figure 1: stage-IV) to established the frequency band reconfigurability feature in proposed design. During forward bias mode (ON-state) of PIN diode, the electric current path length is varied (current perturbation) which derive the hepta (seven) band characteristics in proposed design [38], [39], [40].

During design stage-I (Figure 1), a conventional Octagonal-shaped patch antenna is designed cover the single UWB resonant band. To achieve the multiband characteristics in proposed design, a hybrid fractal approach with metamaterial (SRR cells) loading is implemented. In concern of proposed SRR cell, it is noticed that the perimeter of the conducting split rings (P_{pe − SRL − 1} and P_{pe − SRL − 2}) are related with the length (L_{Int − SRL − 1} and L_{Int − SRL − 2}) and width (W_{Int − SRL − 1} and W_{Int − SRL − 2}) and expressed in following equations:

Above equations are represent the relation between the perimeter of the conducting split rings (P_{pe − SRL − 1} and P_{pe − SRL − 2}) and guided wavelength (λ_guw) at the respective operating/resonant frequency (f_reso). The effective dielectric constant (ε_effec) is related with substrate height (h_subst), feedline width (W_fdl) and relative permittivity of substrate (ε_rp) as indicated in Eq. (3) [38]. As per design evolution stages of proposed design, when introducing the SRR cells in antenna (Design stage-III: Figure 1) an additional resonant band at 4.41 GHz (Lower C band) is achieved as illustrated in Figure 2.

In hybrid fractal metamaterial multiband antenna, multiple modes are excited with application of feedline across the radiating patch with respective to the various resonant bands. The fundamental modes (includes higher order odd modes) and even modes (second and fourth order) are formed across the middle section of slotted hybrid fractal radiating patch area (exhibit high impedance) and SRR cells surface respectively. It is also observable that the odd/even modes are excited along the central radiating part depends upon the application of the high/low impedance source feeding to the proposed structure, respectively. The respective generated even/odd modes are liable to establish the appropriate frequency response for proposed design at various wireless standards.

The characteristics mode theory (CMT) derive the relation between the antenna conductor surface current J_co − sur and characteristic mode currents J_cmc (linear summation formation) as:

where γ_k is the MWC of the kth characteristic mode [41].

Another observation for ideal conductor surface is that the electric field (because of conductor surface current) is equal and opposite manner to the tangential incident electric field is identified as:

Above equation represent the relation between the source and the field. On solving Eq. (5) with the help of Eq. (4) as:

Above expression is modified (with introduction of dot product and parameter J_w (w is any integer)) as:

The characteristic impedance Z_ch−im = R_e + jX_i has real and imaginary terms which are related with radiated and stored energy, respectively.

The orthogonality features is present in characteristic modes and define as [41]:

where,

δwk = Kronecker Delta,

λk = Eigenvalue with respect to mode k, and

Pcomplx(Jw,Jk) = Created complex power.

The value of γ_k (MWC of the kth characteristic mode) is calculated with the help of Eqs. (7) and (8) as:

Above equation may be represented in matrix form as follows:

where,

γk = Modal weight coefficients,

Vex−so−cl = Excitation source column matrix, and

Ich−cl = Characteristic current column matrix of mode k (generated by an eigenvalue equation).

The characteristic current column matrix I_ch − cl of the conductor surface is written as [41]:

The resultant seven (hepta) modes are excited at WiMAX (3.5 GHz)/Lower C-band (4.41 GHz)/WLAN (5.4/5.8 GHz)/Lower X-band (8.26 GHz)/Upper X-band (10.48 GHz)/Lower Ku-band (13.35 GHz)/Middle Ku-band (14.42 GHz) wireless standards with respective to the various eigen values. Since theses modes are not identical to each other but respective modes commonly allotted to accomplish the hepta wireless communication states. In concern of respective excited resonant modes, the impedance matching is attained by implementation of slotted hybrid metamaterial radiating section in proposed design. The input impedance of antenna is computed by considering the electromagnetic fields present between the radiating patch and ground section. Accordingly, each operating state is generalized by the parallel resonant RLC circuit formation as represented in Figure 3. The expression of equivalent input impedance of proposed design is given in following Eq. (12) [42]:

Where,

• (Zinput−imp)equiv = Equivalent input impedance at feed point of the proposed design,

• L∞ = Equivalent Inductance at higher operating resonant bands,

• N = Number of radiating operating modes,

• fr−q = Resonant frequency of the qth mode,

• Qfactor,q = Q-factor of qth resonant mode,

• Rrad,q = Radiation resistance of the qth mode, and

• ωreso = 2πf (resonant frequency in radian/second).

Figure 3:

Equivalent circuit model for proposed hybrid fractal metamaterial multiband antenna.

As per concern with design evolution stage-IV, proposed structure is printed on FR4 dielectric material with electrical size 0.3422λ × 0.2552λ × 0.0186λ (29.5 × 22 × 1.6 mm) at lower resonant frequency of 3.48 GHz exhibit trapezoidal shape microstrip feedline and ground plane, as illustrated in Figure 4. The radiating section of antenna has octagonal shape slot (side length 7.4 mm) with three duplication and seven segments of hybrid fractal geometry (fused form of first iterated Moore and Koch curve) loaded with two rectangular SRR cells, as represented in Figure 4 (a). Proposed metamaterial SRR cell consist two rectangular rings attached commonly at one side and opened at opposite side. The layouts (front, side and back view) of proposed structure is indicated in Figure 3. The optimized parametric values (in mm) of antenna design (Figure 4) are:- L_S (substrate length) = 29.5, W_S (substrate width) = 22, h (substrate height) = 1.6, L_G (ground plane length) = 8, W_GR1 (trapezoidal shape ground plane width: lower side) = 22, W_GR2 (trapezoidal shape ground plane width: upper side) = 10, S_L1 (side length: slotted octagonal shape radiating section–larger) = 8 and S_L2 (side length: slotted octagonal shape radiating section–smaller) = 2.

Figure 4:

Layouts of proposed structure:

(a) Front view,

(b) Side view,

(c) Back view.

The proposed hybrid fractal metamaterial multiband antenna is simulated with the help of simulator CST Microwave Studio (MWS) [43] by using finite element method. After simulation process, the proposed design is fabricated by using PCB prototype machine. The top and bottom side of fabricated prototype is depicted in Figure 5.

Figure 5:

Images of fabricated prototype:

(a) Top view (without diode);

(b) Bottom view (without diode);

(c) Top view (with diode);

(d) Bottom view (with diode).

The DC potential is applied across the metal strip (dimension of 2 × 0.6 mm²) to perform the switching between the biasing states (forward/revers bias) of PIN diode. When the DC potential is applied the PIN diode is in forward bias (ON state); and when the applied potential is absent (zero volt), the diode represent the reverse bias mode (OFF state). The PIN diode, metallic strip and blocking capacitor (100 pF) are arranged in series form and placed between the seventh hybrid fractal geometry and feedline, as illustrated in Figure 4 (a). The blocking capacitor is utilized as an isolator to isolate the RF signal from DC part. In proposed design a beam lead PIN diode model of ALPHA-6355 is implemented to perform the frequency-band reconfigurability characteristics during forward bias (ON state–equivalent to 2.6 Ω resistance) and reverse bias (OFF states–equivalent to 0.081 pF capacitance) when applying the DC potential (0.7 V) [44]. Figure 6 (a) and (b) represents the equivalent circuits modeling of ALPHA-6355 PIN diode during ON and OFF states respectively. This modeling denotes the series form of resistor R_CC and fixed value inductor L_F under ON state (forward bias), whereas during OFF state (reverse bias) it represents the parallel form of resistor R_CC and reactive capacitor C_R, in series with fixed value inductor L_F as depicted in Figure 6 (a) and (b).

Figure 6:

An equivalent circuit of PIN diode during ON/OFF states.

The measurement set-up is establish to obtain the experimental parameters like as radiation patterns, gain and efficiency of proposed antenna with the help of vector network analyzer (VNA) and anechoic chamber as shown in Figure 7. In this setup the far-field radiation pattern of proposed design (AUT–Antenna under test) are achieved by using a horn antenna (reference antenna) with Fraunhofer distance [D_F ≫ 2 (D_max)²/λ₀; D_max–maximum dimensions of AUT and λ₀–free space wavelength] from proposed antenna in anechoic chamber [38]. The measured values of E and H plane patterns are retrieved at different resonant frequencies by implementation of step rotation process (rotation step about 5° from 0 to 360° with the help of rotation table) during co/cross polarization mode. Subsequently, another antenna parameter such as reflection coefficient (S-parameter: S₁₁), gain, radiation efficiency and antenna impedance are also measured by using the same setup.

Figure 7:

Setup (anechoic chamber with VNA) to measure various radiation parameters of proposed antenna.

As per the observation of simulated and measured S-parameters (S₁₁), the proposed structure is resonant on hepta/hexa band under the forward (ON state)/reverse bias (OFF state) modes of PIN diode respectively (electric current path length is increase/reduces during ON/OFF state), as depicted in Figure 8 and Table 2. Proposed antenna exhibit hepta-band characteristics with S₁₁ < −10 dB impedance bandwidth of about 0.95 GHz (2.86–3.81 GHz, 28.48%), 0.51 GHz (4.25–4.76 GHz, 11.32%), 2.61 GHz (5.01–7.62 GHz, 41.33%), 2.4 GHz (7.75–10.15 GHz, 26.82%), 0.73 GHz (10.43–11.16 GHz, 6.76%), 1.14 GHz (12.84–13.98 GHz, 8.50%) and 1.77 GHz (14.12–15.89 GHz, 11.79%) during simulation process, and about 0.71 GHz (3.01–3.72 GHz, 21.09%), 0.44 GHz (4.28–4.72 GHz, 9.78%), 2.37 GHz (5.11–7.48 GHz, 37.65%), 2.27 GHz (7.81–10.08 GHz, 25.38%), 0.7 GHz (10.44–11.14 GHz, 6.49%), 1.02 GHz (12.92–13.94 GHz, 7.59%) and 1.68 GHz (14.14–15.82 GHz, 11.21%) under measurement mode. It is noticed that the frequency band reconfigurability features of proposed antenna for hepta/hexa wireless standards is achieved by using ON/OFF states of PIN diode, as mentioned in Table 2.

Figure 8:

Simulated and measured S₁₁ results during ON/OFF switching states of PIN diode.

2.1 Design and analysis of proposed hybrid fractal geometry

The geometrical structure of proposed radiating hybrid fractal part is obtained by applying the fusion process between the two identical fractal curves (Moore and Koch) as illustrated in Figure 9. The creation of hybrid fractal configuration (fractalization) is depend upon the formation of self-repeating and self-similarity geometrical approach with a constant factor in each iteration stages. The parameters of hybrid fractal structure such as fractal dimension (D_frd) and length (L_frt) are achieved with implementation of Iterative Function System (IFS). The respective parameters are calculated using the following equations [45]:

(13)Dfrd=log(Qsm)/log(npr)

(14)Lfrt=Tch(Qsm/npr)p

where,

• Q_sm = Total segments in hybrid fractal geometry,

• n_pr = Total segment partitioning during iteration stages of hybrid fractal geometry,

• T_ch = Height of fractal curve,

• p = Total iteration stages for hybrid fractal geometry.

Figure 9:

Fusion process for hybrid fractal structure (fractal curves: first iterated Moore /Koch curve).

The IFS approach is applicable to drive the mathematical analysis of proposed fractal structure. The respective approach is implemented by using affine transformation including the scaling/translation/rotation operations represented in following equations [45];

(15)T(x)=CX+g=[C1C2C3C4][x1x2]+[g1g2]

(16)C=1m[Cos θ″−Sin θ″Sin θ″Cos θ″]

where

• C = Initial geometry matrix for fractal structure,

• C₁, C₂, C₃, C₄ = Controlling parameters regarding rotation and scaling operation for fractal geometry,

• x₁, x₂ = Coordinating point of vector X,

• g₁, g₂ = Controlling parameters to performing the translation and linear shift operation for fractal geometry,

• m = Constant scaling parameter,

• θ″ = Angle rotational parameter,

• g = Translation parameter for fractal structure.

The proposed hybrid fractal structure is obtained by applying the Hutchinson operator H (C) on the set of transformations for controlling operator ‘C’ as indicated in following Eq. (17) [45];

(17)H(C)=Un=1NHn(C)

The designing of proposed hybrid fractal structure is accomplished by using the fusion process between the basic Moore and Koch curve, as depicted in Figure 9. During fusion process first iterated Moore/Koch curve are integrated with each other to create the proposed Moore-Koch Hybrid Fractal Antenna (MKHFA) (Figure 4). In proposed hybrid fractal geometry, Moore and Koch curve are working as an initiator and generator respectively, where generator is superimposed on upward/downward segment of the initiator as illustrated in Figure 9 (a)–(c). Another observation is that the overall length of the conducting radiating part (seven sections of hybrid fractal geometry) is increases about 11.12% by fusion of Koch curve with Moore fractal curve [Figure 4 (a)]. Initially, the length of base curve (Moore Curve: single section) is 28.4 mm without integration of Koch curve and when Koch curve is superimposed on Moore curve the length of base curve is increases upto 31.36 mm. The optimized parametric values (in mm) of proposed hybrid fractal geometry (Figure 3) are:- L_MC1 (side length) = 1.15, W_MC1 (upper width) = 3, t_MC1 (thickness) = 0.25, W_g1 (gap width) = 0.5, L_MC2 (side length) = 0.9, W_MC2 (upper width) = 2, t_MC2 (thickness) = 0.2, W_g2 (gap width) = 0.4, L_KC (side length) = 0.75, W_KC (width) = 0.8, t_KC (thickness) = 0.25 and W_g3 (gap width) = 0.3.

2.2 Design and analysis of proposed metamaterial SRR

This part of paper mentioned the designing and analysis (simulated and measurement mode) of proposed rectangular metamaterial SRR cell. The SRR geometry has two rectangular shape conducting rings with split gap, as illustrated in Figure 10. The S-parameter analysis of proposed SRR cell is achieve by using the waveguide setup in simulation environment (CST Microwave Studio (MWS) simulator) [43] to obtain the simulated reflection parameter (S₁₁: at port 1) and transmission parameter (S₂₁: at port 2) (Figure 10).

Figure 10:

Waveguide setup for proposed SRR (simulation environment) with equivalent circuit.

During analysis of SRR cell inside the waveguide setup, implement the waveguide port at the side wall instead of front of the cell (known as the magnetic/electric boundaries or unit-cell boundary condition). As per concern with boundary condition, the waveguide port 1/2, PMC and PEC are allocated for X, Y and Z directions respectively for proposed SRR cell analysis in waveguide setup under simulation mode to retrieve the S-parameters (S₁₁ and S₂₁). The generated electric field (E) is orthogonal with respect to the created magnetic field (H) across the SRR cell within waveguide setup. Another observation is that the magnetic field lines are perpendicular to the SRR plane. The electric field (E) is in parallel and perpendicular direction along the magnetic and electric wall respectively and present across the y-axis at the centre of the waveguide. When applying the excited wave from the source at port 1, the excitation modes are created and present inside the waveguide, and plot the S-parameters (S₁₁ and S₂₁) curves (Simulation mode: with the help of simulator, Experimental mode: with the help of VNA).

The effective permeability μ_eff/permittivity ε_eff of proposed SRR cell are retrieved by using the S-parameters S₁₁ and S₂₁ (from the waveguide setup). The measured S-parameters (S₁₁/S₂₁) of proposed SRR are realized with the help of experimental waveguide setup in which the SRR structure is kept inside the waveguide environment attached with two coaxial-to-waveguide adapters (at both ends of waveguide) and VNA (via connecting cable) [46]. The optimized dimensions (in mm) of proposed SRR structure (Figure 4) are:- L_SRRE (length of external rectangular SRR ring) = 2.75, W_SRRE (width of external rectangular SRR ring) = 8, L_SRRI (length of internal rectangular SRR ring) = 2, W_SRRI (width of internal rectangular SRR ring) = 6, t_SRR (thickness of SRR ring) = 0.25, g (gap width) = 1.4, L_SL1 (length of rectangular strip) = 0.75 and W_SL1 (width of rectangular strip) = 0.5.

In waveguide setup, during the application of magnetic field on the proposed SRR, an EMF (electro-magnetic field) is created and present across the SRR structure. The generated EMF is coupled with the two conducting rings (rectangular shape) of SRR cell. In this state, conduction current is passing from outer to inner rectangular ring through a distributed capacitance (Capacitive effect: because of ring spacing/gap ‘g’). Therefore, the respective formation (SRR structure) is generating the LC resonant characteristics and provide a specific operating frequency calculated by using the following Eq. (18) [47], [48], [49];

(18)fr−rssr= 12π(Ltot−eqt)(Ctot−eqt)  

where

(19)Ltot−eqt=0.0002[{2(LExt−rsrr+WExt−rsrr)−g)}{loge8(LExt−rsrr+WExt−rsrr−0.5g)/t}−δ],

(20)Ctot−eqt=0.5[{(LExt−rsrr+Wext−rsrr)−π(t+0.5g)−g}{(0.3×10−8)(SQRT(εe))/Z0}+{(8.85×10−12)ht/g}],

• L_{tot − eqt} = Total equivalent inductance of proposed rectangular SRR,

• C_{tot − eqt} = Total equivalent capacitance of proposed rectangular SRR,

• L_{Ext − rsrr} = Length of external rectangular metallic ring of proposed SRR,

• W_{Ext − rsrr} = Width of external rectangular metallic ring of proposed SRR,

• (g₂)_rsrr = (g₁)_rsrr = (g)_rsrr = g = Split gap of internal/external rectangular metallic rings of SRR,

• (t₂)_{Ext − rsrr} = (t₁)_{Ext − rsrr} = (t)_srr = t = Thickness of internal/external rectangular metallic rings of SRR, δ = Constant parameter (with fixed value = 2.451),

• ε_e = Effective dielectric constant (permittivity) of the medium,

• Z₀ = Characteristic impedance,

• h = height of the conducting strip.

The effective dielectric parameter (permeability: μ_{eff − rsrr}) of proposed rectangular SRR can be evaluated by using following Eq. (21) [47], [50];

(21)μeff−rsrr=nrefr−ind×zimpd

where

(22)Refractive index (nrefr−ind)=(1/kwanTslt)cos−1[(1/2S21){1−(S11)2+(S21)2}]

(23)Impedance(zimpad)=[{(1+S11)2−(S21)2}/{(1−S11)2−(S21)2]1/2

Wave number (k_wan) = 2πf/(3 × 10⁸ m/s)

Slab (dielectric substrate material) thickness = T_slt

The comparison between simulated and measured S-parameters (S₁₁: reflection parameter, S₂₁: transmission parameter) of proposed SRR (retrieved from simulated/experimental waveguide setup), as represented in Figure 11. The transmission peak is observed at resonant frequency 4.41 GHz (Figure 11), which can be verified from the theoretical value calculated with the help of Eq. (6). Proposed SRR geometry is identified as a magnetically resonator represents the negative permeability characteristics at operating frequency 4.41 GHz due to the induced magnetic field in perpendicular mode. It is also noticed that the reflection S-parameter (S₁₁) and transmission S-parameter (S₂₁) has about zero level (<−2 dB) and below reference level (−10 dB) respectively at operating frequency 4.41 GHz, which witnesses the stop band features of proposed SRR at this respective resonant frequency. This stop band nature of proposed SRR represent the negative permeability characteristics at resonant frequency 4.41 GHz, as depicted in Figure 12.

Figure 11:

Comparison of simulated and measured S₁₁/S₂₁-parameter (retrieved from waveguide setup).

Figure 12:

Simulated and measured permeability (real and imaginary mue) curve of proposed SRR.

3 Results

The working mechanism of proposed hybrid fractal metamaterial multiband antenna can be easily define by analysis of current distribution plots at resonant modes 3.5 GHz (WiMAX)/4.41 GHz (Lower C-band)/5.4/5.8 GHz (WLAN)/8.26 GHz (Lower X-band)/10.48 GHz (Upper X-band)/13.35 GHz (Lower Ku-band)/14.42 GHz (Middle Ku-band) as illustrated in Figure 13. At lower resonant frequency mode (wireless standard: WiMAX–3.5 GHz), maximum current distribution is allocated along the periphery of centered seventh hybrid (Moore/Koch) fractal geometry (attached with feedline via strip) as shown in Figure 13. For higher operating frequency mode (wireless standard: Lower C band–4.41 GHz), the concentration of current is maximally focused across the surface of the two SRR cells. Next for higher resonant frequency modes (wireless standard: WLAN–5.4/5.8 GHz, Lower X band–8.26 GHz and Middle Ku band–14.42 GHz), the maximum surface current distribution is exist along the surface of the six hybrid fractal sections (three duplications) in radiating part, which provides the enhanced operating bandwidth with better impedance matching. From Figure 13, it is also noticed that the surface current is exceedingly circulated across the surface of the seventh centered hybrid fractal geometry (connected with feedline via conducting strip) and inner section of octagonal shape slotted radiating section (periphery of six octagonal shape part with hybrid fractal structures, and surface of inner rectangular conducting ring of SRR cells) for higher operating frequency modes 10.48 GHz (upper X band) and 13.35 GHz (Lower Ku band) respectively. Another observation is that the surface current distribution is minimum across the edges of SRR cells/hybrid fractal geometries, and maximum spreads out along the surfaces of various parts (seven hybrid fractal structures/two SRR cells) of radiating patch for all the resonant frequency wireless communication modes, provides the enhanced form of radiation characteristics and bandwidth with improved impedance matching.

Figure 13:

Current distribution analysis (during simulation process: PIN Diode–ON State [48], [49]) for proposed design at various resonant frequency wireless communication modes.

From simulated and measured gain plot of proposed antenna structure, peak gain of 1.98, 2.38, 2.62, 3.91, 2.24, 3.28 and 4.11 dBi (in simulation process), and 1.72, 1.74, 1.98, 2.68, 2.01, 2.98 and 3.79 dBi (during experimental process) are observed at seven (hepta) wireless communication resonant modes 3.5 GHz (WiMAX), 4.41 GHz (Lower C-band), 5.8 GHz (WLAN), 8.26 GHz (Lower X-band), 10.48 GHz (Upper X-band), 13.35 GHz (Lower Ku-band) and 14.42 GHz (Middle Ku-band) respectively as represented in Figure 14.

Figure 14:

Simulated and experimental gain plot of projected multiband antenna.

The comparison of simulated and measured radiation efficiency of projected hybrid fractal metamaterial multiband antenna is analyzed in Figure 15. Proposed structure represent the radiation efficiencies 51.25, 54.28, 59.69, 67.82, 72.34, 79.92 and 81.64% (under simulation process) and 44.56, 48.56, 52.76, 59.89, 71.04, 74.89 and 79.12% (during experimental process) at multiple wireless communication resonant bands 3.5 GHz (WiMAX), 4.41 GHz (Lower C-band), 5.8 GHz (WLAN), 8.26 GHz (Lower X-band), 10.48 GHz (Upper X-band), 13.35 GHz (Lower Ku-band) and 14.42 GHz (Middle Ku-band) respectively as depicted in Figure 15.

Figure 15:

Comparison of simulated and experimental radiation efficiency against operating frequency range.

To enhance the study about the radiation characteristics of proposed antenna another parameter E & H plane patterns (two dimensional radiation patterns) are analyzed in this section. The principle plane (E & H) patterns (co/cross-polarization states) are observed during simulation and measurement modes, showing a better agreement between these two mode results. The patterns are plotted at distinct operating wireless communication frequencies 3.5 GHz (WiMAX), 4.41 GHz (Lower C-band), 5.8 GHz (WLAN), 8.26 GHz (Lower X-band), 10.48 GHz (Upper X-band), 13.35 GHz (Lower Ku-band) and 14.42 GHz (Middle Ku-band) respectively as indicated in Figure 16. It is observed that, the patterns are arbitrary bidirectional/dipole like and omnidirectional in case of E and H plane (under co-polarization state) respectively, which validate the proposed antenna is most suitable for wireless communication applications. The patterns are achieve the low level of cross polarization (below −15 dB) with stability and consistency at all the operating frequencies.

Figure 16:

Two dimensional patterns (E and H plane: co/cross-polarization states) of proposed antenna at 3.5, 4.41, 5.8, 8.26, 10.48, 13.35 and 14.42 GHz.

Table 1 illustrates the performance comparison conducted between the proposed antenna and the existing reported multiband antennas. The projected antenna exhibits seven wireless communication operating bands (with frequency band reconfigurability characteristics), compact dimensions, enhanced bandwidth, acceptable gain and efficiency.

4 Conclusion

In this research article, a hybrid fractal metamaterial loaded hepta band antenna is designed to cover the S band WiMAX (3.5 GHz), Lower C band (4.41 GHz: terrestrial fixed and mobile broadband application), C band WLAN (5.4/5.8 GHz), Lower X (8.26 GHz: band Earth exploration-satellite service ITU region 2), Upper X band (10.48 GHz: Amateur satellite operating band), Lower Ku band (13.35 GHz: Radar communication application) and Middle Ku band (14.42 GHz: Geostationary satellite service) wireless communication applications. The proposed structure exhibit slotted radiating part with seven segments (three duplication sections) of hybrid fractal (Moore-Koch curve) geometry and two rectangular metamaterial SRR cells to achieve the multiple resonant characteristics. Antenna structure also present the frequency band reconfigurability features to obtain the band switching between the wireless standards by implementation of PIN diode within radiating patch. Proposed design has stable and consistent E & H plane patterns (low level of cross polarization), enhanced bandwidth, acceptable gain and efficiency at all the hepta (seven) operating bands of wireless standards. For future work, hybrid fractal approach with multiple fractal curve (2–3 curve) with higher iteration states can be applied to achieve multiband (more than hepta resonant bands) characteristics in the proposed antenna. Since the proposed design has maximum peak gain about 4.11 dBi at 14.42 GHz (Middle Ku-band) during simulation, so the gain can be enhanced by applying the approach of FSS implementation with antenna design.